MOTOROLA Order this document SEMICONDUCTOR APPLICATION NOTE by AN1541/D AN1541 Introduction to Insulated Gate Bipolar Transistors Prepared by: Jack Takesuye and scott Deuty Motorola inc INTRODUCTION As power conversion relies more on switched applications The IGBT is, in fact, a spin-off from power MOSFET semiconductor manufacturers need to create products that technology and the structure of an iGBt closely resembles approach the ideal switch. The ideal switch would have: that of a power MOSFET. The IGBT has high input impedance 1)zero resistance or forward voltage drop in the on-state, and fast turn-on speed like a MOSFET IGB Ts exhibit an 2)infinite resistance in the off-state, 3)switch with infinite on-voltage and current density comparable to a bipolar speed, and 4)would not require any input power to make it transistor while switching much faster. IGBTs are replacing MOSFETs in high voltage applications where conduction When using existing solid-state switch technologies, the losses must be kept low. With zero current switching or designer must deviate from the ideal switch and choose a resonant switching techniques, the IGBT can be operated in device that best suits the application with a minimal loss of the hundreds of kilohertz range [1] efficiency. The choice involves considerations such as Although turn-on speeds are very fast, turn-off of the IGBT voltage, current, switching speed, drive circuitry, load, and is slower than a MOSFET. The IGBT exhibits a current fall time temperature effects. There are a variety of solid state switch or tailing The tailing restricts the devices to operating at technologies available to perform switching functions moderate frequencies(less than 50 kHz)in traditional"square however, all have strong and weak points. waveform"PWM, switching applications HIGH VOLTAGE POWER MOSFETS At operating frequencies between 1 and 50 kHZ, IGB Ts offe an attractive solution over the traditional bipolar transistor The primary characteristics that are most desirable in a MOSFETS and thyristors. Compared to thyristors, the IGBT is solid-state switch are fast switching speed, simple drive faster, has better dv/dt immunity and, above all, has better gate requirements and low conduction loss. For low voltage turn-off capability. While some thyristors such as GTOs are applications, power MOSFETs offer extremely low capable of being turned off at the gate, substantial reverse on-resistance, RDS(on), and approach the desired ideal gate current is required, whereas turning off an IGBT only switch. In high voltage applications, MOSFETs exhibit requires that the gate capacitance be discharged. A thyristor increased RDS(on) resulting in lower efficiency due to has a slightly lower forward-on voltage and higher surge capability than an IGBT. h-resistance is proportional to the breakdown voltage raised MOSFETs are often used because of their simple gate drive to approximately the 2.7 power(1) requirements. Since the structure of both devices are so MOSFET technology has advanced to a point similar, the change to IGB Ts can be made without having to densities are limited by manufacturing equipment capabilities redesign the gate drive circuit. IGBTS, like MOSFETS, are and geometries have been optimized to a point where the transconductance devices and can remain fully on by keeping g DS(on)is near the predicted theoretical limit. Since the cell the gate voltage above a certain threshold As shown in Figure 1a, using an IGBT in place of a play a major role, no significant reduction in the RDS(on)is MOSFET dramatically reduces the forward voltage drop at els above 12 amps. By reducing the forward drop the conduction loss of the device is decreased. The gradua problem of increased on-resistance without sacrificing rising slope of the MOSFET in Figure la can be attributed to witching speed the relationship of VDs to RDS(on ). The IGBT curve has ar offset due to an internal forward biased p-n junction and a fast DSS rising slope typical of a minority carrier device It is possible to replace the MOSFET with an IGBT and improve the efficiency and/or reduce the cost. As shown in ENTER THE IGBT Figure 1b, an iGBT has considerably less silicon area than a similarly rated MOSFET Device cost is related to silicon area; By combining the low conduction loss of a BjT with the therefore, the reduced silicon area makes the IGBT the lower witching speed of a power MOSFET an optimal solid state cost solution. Figure 1c shows the resulting package area switch would exist. The Insulated-Gate Bipolar Transistor reduction realized by using the IGBT. The IGBT is more space (IGBT)technology offers a combination of these attributes efficient than an equivalently rated MOSFET which makes it MOOROLA e Motorola Inc. 1995
MOTOROLA 1 Prepared by: Jack Takesuye and Scott Deuty Motorola Inc. INTRODUCTION As power conversion relies more on switched applications, semiconductor manufacturers need to create products that approach the ideal switch. The ideal switch would have: 1) zero resistance or forward voltage drop in the on–state, 2) infinite resistance in the off–state, 3) switch with infinite speed, and 4) would not require any input power to make it switch. When using existing solid–state switch technologies, the designer must deviate from the ideal switch and choose a device that best suits the application with a minimal loss of efficiency. The choice involves considerations such as voltage, current, switching speed, drive circuitry, load, and temperature effects. There are a variety of solid state switch technologies available to perform switching functions; however, all have strong and weak points. HIGH VOLTAGE POWER MOSFETs The primary characteristics that are most desirable in a solid–state switch are fast switching speed, simple drive requirements and low conduction loss. For low voltage applications, power MOSFETs offer extremely low on–resistance, RDS(on), and approach the desired ideal switch. In high voltage applications, MOSFETs exhibit increased RDS(on) resulting in lower efficiency due to increased conduction losses. In a power MOSFET, the on–resistance is proportional to the breakdown voltage raised to approximately the 2.7 power (1). MOSFET technology has advanced to a point where cell densities are limited by manufacturing equipment capabilities and geometries have been optimized to a point where the RDS(on) is near the predicted theoretical limit. Since the cell density, geometry and the resistivity of the device structure play a major role, no significant reduction in the RDS(on) is foreseen. New technologies are needed to circumvent the problem of increased on–resistance without sacrificing switching speed. RDS(on) V 2.7 DSS (1) ENTER THE IGBT By combining the low conduction loss of a BJT with the switching speed of a power MOSFET an optimal solid state switch would exist. The Insulated–Gate Bipolar Transistor (IGBT) technology offers a combination of these attributes. The IGBT is, in fact, a spin–off from power MOSFET technology and the structure of an IGBT closely resembles that of a power MOSFET. The IGBT has high input impedance and fast turn–on speed like a MOSFET. IGBTs exhibit an on–voltage and current density comparable to a bipolar transistor while switching much faster. IGBTs are replacing MOSFETs in high voltage applications where conduction losses must be kept low. With zero current switching or resonant switching techniques, the IGBT can be operated in the hundreds of kilohertz range [1]. Although turn–on speeds are very fast, turn–off of the IGBT is slower than a MOSFET. The IGBT exhibits a current fall time or “tailing.” The tailing restricts the devices to operating at moderate frequencies (less than 50 kHz) in traditional “square waveform” PWM, switching applications. At operating frequencies between 1 and 50 kHz, IGBTs offer an attractive solution over the traditional bipolar transistors, MOSFETs and thyristors. Compared to thyristors, the IGBT is faster, has better dv/dt immunity and, above all, has better gate turn–off capability. While some thyristors such as GTOs are capable of being turned off at the gate, substantial reverse gate current is required, whereas turning off an IGBT only requires that the gate capacitance be discharged. A thyristor has a slightly lower forward–on voltage and higher surge capability than an IGBT. MOSFETs are often used because of their simple gate drive requirements. Since the structure of both devices are so similar, the change to IGBTs can be made without having to redesign the gate drive circuit. IGBTs, like MOSFETs, are transconductance devices and can remain fully on by keeping the gate voltage above a certain threshold. As shown in Figure 1a, using an IGBT in place of a power MOSFET dramatically reduces the forward voltage drop at current levels above 12 amps. By reducing the forward drop, the conduction loss of the device is decreased. The gradual rising slope of the MOSFET in Figure 1a can be attributed to the relationship of VDS to RDS(on). The IGBT curve has an offset due to an internal forward biased p–n junction and a fast rising slope typical of a minority carrier device. It is possible to replace the MOSFET with an IGBT and improve the efficiency and/or reduce the cost. As shown in Figure 1b, an IGBT has considerably less silicon area than a similarly rated MOSFET. Device cost is related to silicon area; therefore, the reduced silicon area makes the IGBT the lower cost solution. Figure 1c shows the resulting package area reduction realized by using the IGBT. The IGBT is more space efficient than an equivalently rated MOSFET which makes it perfect for space conscious designs. Order this document by AN1541/D SEMICONDUCTOR APPLICATION NOTE Motorola, Inc. 1995
AN1541 0.10 35 N60D IGBT (0.17X0227) A MOSFET DIE SIZE (0.35X0.26) MOSFET Figure 1b. Reduced Die Size of IGBT Realized When Compared to a MOSFET with Similar Ratings FORWARD DROP (VOLTS) Figure 1a. Reduced Forward Voltage Drop of IGBT Realized When Compared to a MOSFET with Similar Ratings SIZE (TO- 若0 MOSFETPACKAGE terms of voltage and current. However, the presence of an isolated gate in an IGBT makes it simpler to drive than a BJT. BJTs require that base current be continuously supplied in a quantity sufficient enough to maintain saturation. Base currents of one-tenth of the collector current are typical to keep a Bjt in saturation. bjt drive circuits must be sensitive to variable load conditions the base current of a bjt must be kept proportional to the collector current to prever Figure 1c. Reduced Package Size of IGBT Realized desaturation under high-current loads and excessive base When Compared to a MOSFET with Similar Ratings drive under low-load conditions This additional base current increases the power dissipation of the drive circuit. BJTs are Because the loss period is a small percentage of the total on minority carrier devices and charge storage effects including time, slower switching is traded for lower conduction loss. In recombination slow the performance when compared to a higher frequency application, just the opposite would be true majority carrier devices such as MOSFETS. IGBTs also and the device would be made faster and have greater experience recombination that accounts for the current conduction losses. Notice that the curves in Figure 2 show ailing yet IGBTs have been observed to switch faster than reductions in both the forward drop(VCE(sat)and the fall time, tf of vices Thus far, the igbT has demonstrated certain advantages IGBT the device of choice for applications such as motor over power MOSFETs with the exception of switching spee drives, power supplies and inverters that require devices rated Since the initial introduction of IGBTS in the early 1980s for 600 to 1200 volts semiconductor manufacturers have leamed how to make the devices faster. As illustrated in Figure 2, some trade-offs in conduction loss versus switching speed exist. Lower frequency applications can tolerate slower switching devices 3.5 3.0 1ST GENERATION COMPETITOR ND GENERATION COMPETITOR 3RD GENERATION COMPETITOR 1993 OW SATURATION SERIES SERIES DEMONSTRATED Technologies for Forward Voltage Drop (VCE(sat)and Fall Time (tf) MOTOROLA
2 MOTOROLA FORWARD DROP (VOLTS) Figure 1a. Reduced Forward Voltage Drop of IGBT Realized When Compared to a MOSFET with Similar Ratings 40 35 30 25 20 15 10 5 0 0 2 4 6 8 10 PEAK CURRENT THROUGH DEVICE (AMPS) VCE(sat) MGW20N60D IGBT VDS MTW20N50E MOSFET When compared to BJTs, IGBTs have similar ratings in terms of voltage and current. However, the presence of an isolated gate in an IGBT makes it simpler to drive than a BJT. BJTs require that base current be continuously supplied in a quantity sufficient enough to maintain saturation. Base currents of one–tenth of the collector current are typical to keep a BJT in saturation. BJT drive circuits must be sensitive to variable load conditions. The base current of a BJT must be kept proportional to the collector current to prevent desaturation under high–current loads and excessive base drive under low–load conditions. This additional base current increases the power dissipation of the drive circuit. BJTs are minority carrier devices and charge storage effects including recombination slow the performance when compared to majority carrier devices such as MOSFETs. IGBTs also experience recombination that accounts for the current “tailing” yet IGBTs have been observed to switch faster than BJTs. Thus far, the IGBT has demonstrated certain advantages over power MOSFETs with the exception of switching speed. Since the initial introduction of IGBTs in the early 1980s, semiconductor manufacturers have learned how to make the devices faster. As illustrated in Figure 2, some trade–offs in conduction loss versus switching speed exist. Lower frequency applications can tolerate slower switching devices. ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÇÇÇÇÇÇ ÇÇÇÇÇÇ ÇÇÇÇÇÇ ÇÇÇÇÇÇ Figure 1b. Reduced Die Size of IGBT Realized When Compared to a MOSFET with Similar Ratings 0.10 0.05 0 1 AREA (SQ. INCHES) IGBT DIE SIZE (0.17 X 0.227) MOSFET DIE SIZE (0.35 X 0.26) Ç Ç É ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÉÉÉÉÉÉ ÇÇÇÇÇÇ ÇÇÇÇÇÇ ÇÇÇÇÇÇ ÇÇÇÇÇÇ Figure 1c. Reduced Package Size of IGBT Realized When Compared to a MOSFET with Similar Ratings 0.60 0.20 0 1 AREA (SQ. INCHES) IGBT PACKAGE SIZE (TO–220) MOSFET PACKAGE SIZE (T0–247) Ç Ç É 0.40 Because the loss period is a small percentage of the total on time, slower switching is traded for lower conduction loss. In a higher frequency application, just the opposite would be true and the device would be made faster and have greater conduction losses. Notice that the curves in Figure 2 show reductions in both the forward drop (VCE(sat)) and the fall time, tf of newer generation devices. These capabilities make the IGBT the device of choice for applications such as motor drives, power supplies and inverters that require devices rated for 600 to 1200 volts. Figure 2. Advanced Features Offered by the Latest Motorola IGBT Technologies for Forward Voltage Drop (VCE(sat)) and Fall Time (tf) tf (µs) 3.5 3.0 2.5 2.0 1.5 1.0 0.5 00 0.9 0.2 0.3 0.4 0.5 0.6 0.7 0.8 1.0 1ST GENERATION COMPETITOR 1985 2ND GENERATION COMPETITOR 1989 1ST GENERATION MOTOROLA 1993 3RD GENERATION COMPETITOR 1993 2ND GENERATION MOTOROLA DEMONSTRATED VCE(sat) (VOLTS) HIGH SPEED SERIES LOW SATURATION SERIES 0.1
AN1541 CHARACTERISTICS OF IGBTS: DEVICE STRUCTURE The structure of an igbt is similar to that of a double The n--epi resistivity determines the breakdown voltage of diffused(DMOS)power MOSFET. One difference between a MOSFET as mentioned earlier using relationship(1) MOSFET and an IGBT is the substrate of the starting material By varying the starting material and altering certain process steps, an iGBT may be produced from a power MOSFET mask; however, at Motorola mask sets are designed To increase the breakdown voltage of the MOSFET, the specifically for IGBTs. In a moSfET the substrate is N+ as n-epi region thickness (vertical direction in figure)is shown in Figure 3b. the substrate for an iGBT is P+ as shown increased. As depicted in the classical resistance relationship in Figure 3a (2), reducing the RDS(on) of a high voltage device requires greater silicon area a to make up for the increased n-epi EMITTER ■ METAL 口so2 Device designers were challenged to overcome the effects POLYSILICON GATE of the high resistive n- epi region. The solution to this came in the form of conductivity modulation. The n- epi region to this was placed on the Pt substrate forming a p-n junction where conductivity modulation takes place. Because of conductivity modulation, the IGBT has a much greater current density thai a power MOSFET and the forward voltage drop is reduced Now the P+ substrate, n-epi layer and P+"emitter"forma BJT transistor and the n-epi acts as a wide base region N-EPI The subject of current tailing has been mentioned several N+ BUFFER times. Thus far, the device structure as shown in Figure 3 provides insight as to what causes the tailing. Minority carriers P+ SUBSTRATE build up to form the basis for conductivity modulation. When the device turns off, these carriers do not have a current path to exit the device. Recombination is the only way to eliminate the stored charge resulting from the build-up of excess carriers. Additional recombination centers are formed by placing an N+ buffer layer between the n-epi and P+ Figure 3a. Cross Section and Equivalent Schematic substrate of an Insulated Gate Bipolar Transistor(IGBT) Cell While the N+ buffer layer may speed up the recombination it also increases the forward drop of the device. Hence the tradeoff between switching speed becomes a factor in optimizing device performance Additional benefits of the N+ buffer layer include preventing SOURCE thermal runaway and punch-through of the depletion region ■ METAL This allows a thinner n-epi to be used which somewhat POLYSILICON GATE decreases forward voltage drop JFET COLLECTOR GATE一 shorted Figure 4a. IGBT Schematic Symbol N+ SUBSTRATE Figure 3b. Cross Section and Equivalent Schematic of an metalOxide-Semiconductor Field-Effect Transistor (MOSFET Cell Figure 4b. MOSFET Schematic Symbol MOTOROLA
MOTOROLA 3 CHARACTERISTICS OF IGBTs: DEVICE STRUCTURE The structure of an IGBT is similar to that of a double diffused (DMOS) power MOSFET. One difference between a MOSFET and an IGBT is the substrate of the starting material. By varying the starting material and altering certain process steps, an IGBT may be produced from a power MOSFET mask; however, at Motorola mask sets are designed specifically for IGBTs. In a MOSFET the substrate is N+ as shown in Figure 3b. The substrate for an IGBT is P+ as shown in Figure 3a. Figure 3a. Cross Section and Equivalent Schematic of an Insulated Gate Bipolar Transistor (IGBT) Cell POLYSILICON GATE N+ P+ N– EPI N+ BUFFER P+ SUBSTRATE P– Rmod EMITTER N+ P+ P– COLLECTOR GATE NPN MOSFET PNP KEY METAL SiO2 Rshorting Figure 3b. Cross Section and Equivalent Schematic of an Metal–Oxide–Semiconductor Field–Effect Transistor (MOSFET) Cell JFET channel Drain–to–Source Body Diode (Created when NPN base–emitter is properly shorted by source metal) POLYSILICON GATE N+ P+ N– EPI N+ SUBSTRATE SOURCE N+ P+ DRAIN NPN KEY METAL SiO2 GATE P– The n– epi resistivity determines the breakdown voltage of a MOSFET as mentioned earlier using relationship (1). RDS(on) V 2.7 DSS (1) To increase the breakdown voltage of the MOSFET, the n– epi region thickness (vertical direction in figure) is increased. As depicted in the classical resistance relationship (2), reducing the RDS(on) of a high voltage device requires greater silicon area A to make up for the increased n– epi region. R 1 A (2) Device designers were challenged to overcome the effects of the high resistive n– epi region. The solution to this came in the form of conductivity modulation. The n– epi region to this was placed on the P+ substrate forming a p–n junction where conductivity modulation takes place. Because of conductivity modulation, the IGBT has a much greater current density than a power MOSFET and the forward voltage drop is reduced. Now the P+ substrate, n– epi layer and P+ “emitter” form a BJT transistor and the n– epi acts as a wide base region. The subject of current tailing has been mentioned several times. Thus far, the device structure as shown in Figure 3 provides insight as to what causes the tailing. Minority carriers build up to form the basis for conductivity modulation. When the device turns off, these carriers do not have a current path to exit the device. Recombination is the only way to eliminate the stored charge resulting from the build–up of excess carriers. Additional recombination centers are formed by placing an N+ buffer layer between the n– epi and P+ substrate. While the N+ buffer layer may speed up the recombination, it also increases the forward drop of the device. Hence the tradeoff between switching speed and conduction loss becomes a factor in optimizing device performance. Additional benefits of the N+ buffer layer include preventing thermal runaway and punch–through of the depletion region. This allows a thinner n– epi to be used which somewhat decreases forward voltage drop. Figure 4b. MOSFET Schematic Symbol Figure 4a. IGBT Schematic Symbol COLLECTOR EMITTER GATE GATE SOURCE DRAIN
AN154 The IGBT has a four layer(P-N-P-N)structure. This The IGBTs on-voltage is represented by sum of the offset structure resembles that of a thyristor device known as a voltage of the collector to base junction of the PNP transistor licon Controlled Rectifier(SCR). Unlike the SCR where the the voltage drop across the modulated resistance Rmod and device latches and gate control is lost, an IGBT is designed the channel resistance of the intemal MOSFET. Unlike the so that it does not latch on. full control of the device can be MOSFET where increased temperature results in increased maintained through the gate drive RDS(on)and increased forward voltage drop, the forward To maximize the performance of the IGBT, process steps drop of an IGBT stays relatively unchanged at increased e optimized to control the geometry, doping and lifetime he possibility of latching is also reduced by strategic processing of the device. Geometry and doping levels are Switching Speed optimized to minimize the on-voltage, switching speed and Until recently, the feature that limited the IGBT from achieve other key parametric variations. Because the IGBT is serving a wide variety of applications was its relatively slow a four-layer structure, it does not have the inverse parallel turn-off speed when compared to a power MOSFET.While diode inherent to power MOSFETs. This is a disadvantage to turn-on is fairly rapid, initial IGBTs had current fall times of motor control designers who use the anti-parallel diode to around three microseconds recover energy from the motor The turn-off time of an IGBT is slow because many minority Like a power MOSFET, the gate of the IGBT is electrically carriers are stored in the n-epi region When the gate is isolated from the rest of the chip by a thin layer of silicon initially brought below the threshold voltage, the nepi dioxide, SIO2. The IGBT has a high input impedance due to contains a very large concentration of electrons and there will the isolated gate and it exhibits the accompanying be significant injection into the P+ substrate and a dvantages of modest gate drive requirements and excellent corresponding hole injection into the n-epi As the electron gate drive efficiency concentration in the n-region decreases, the electron njection decreases, leaving the rest of the electrons to Equivalent Circuit of IGBT recombine. Therefore. the turn -off of an igbt has two Figure 4b shows the terminals of the IGBT as determined by phases: an injection phase where the collector current falls JEDEC. Notice that the iGBT has a gate like a MOSFET yet very quickly, and a recombination phase in which the collector it has an emitter and a collector like a bjt. current decrease more slowly. Figure 5 shows the switching referring to the cross section of the device and its equivalent designed for PWM motor control seve ors of afast"IGBT The operation of the IGBT is best understood by again waveform and the tail time contributing fa circuit as shown in Figure 3a. Current flowing from collector to emitter must pass through a p-n junction formed by the P+ substrate and n-epi layer. This drop is similar to that seen in TAIL TIME of MOTOROLA GEN. 2 IGBT #2 in a forward biased p-n junction diode and results in an offset 1.0 hp MOTOR DRIVE at 1750 RPM voltage in the output characteristic. Current flow contributions are shown in Figure 3a using varying line thickness with the thicker lines indicating a high current path. For a fast device the N+ buffer layer is highly doped for recombination and MOSFET TURN-OFF speedy turn off. The additional doping keeps the gain of the PORTION PNP low and allows two-thirds of the current to flow through 3 the base of the PNP(electron current )while one-third passes PNP TURN-OFF PORTION through the collector(hole current) Shorting is the parasiticresistance of the Pt emitter region Current flowing through Shorting can result in a voltage across the base-emitter junction of the NPN. If the base-emitter voltage is above a certain threshold level, the NPN will begin to conduct causing the NPN and PNP to enhance each others current flow and both devices can become saturated. This results in the device latching in a Figure 5. IGBT Current Turn-off Waveform fashion similar to an SCR Device processing directs currents withinthe device and keeps the voltage across Shorting lowto In power MOSFETS, the switching speed can be greatly avoid latching. The IGBT can be gated off unlike the SCr affected by the impedance in the gate drive circuit. Efforts to which has to wait for the current to cease allowing minimize gate drive impedance for IGBTs are also recombination to take place in order to turn off IGBTs offer an recommended. Also, choose an optimal device based on advantage over the Scr by controlling the current with the device, not the device with the current. The internal MOSFET switching speed or use a slower device with lower forward of the iGBT when gated off will stop current flow and at that turn-off mechanism is suggested in a paper by Baliga et al [2] point, the stored charges can only be dissipated through MOTOROLA
4 MOTOROLA The IGBT has a four layer (P–N–P–N) structure. This structure resembles that of a thyristor device known as a Silicon Controlled Rectifier (SCR). Unlike the SCR where the device latches and gate control is lost, an IGBT is designed so that it does not latch on. Full control of the device can be maintained through the gate drive. To maximize the performance of the IGBT, process steps are optimized to control the geometry, doping and lifetime. The possibility of latching is also reduced by strategic processing of the device. Geometry and doping levels are optimized to minimize the on–voltage, switching speed and achieve other key parametric variations. Because the IGBT is a four–layer structure, it does not have the inverse parallel diode inherent to power MOSFETs. This is a disadvantage to motor control designers who use the anti–parallel diode to recover energy from the motor. Like a power MOSFET, the gate of the IGBT is electrically isolated from the rest of the chip by a thin layer of silicon dioxide, SiO2. The IGBT has a high input impedance due to the isolated gate and it exhibits the accompanying advantages of modest gate drive requirements and excellent gate drive efficiency. Equivalent Circuit of IGBT Figure 4b shows the terminals of the IGBT as determined by JEDEC. Notice that the IGBT has a gate like a MOSFET yet it has an emitter and a collector like a BJT. The operation of the IGBT is best understood by again referring to the cross section of the device and its equivalent circuit as shown in Figure 3a. Current flowing from collector to emitter must pass through a p–n junction formed by the P+ substrate and n– epi layer. This drop is similar to that seen in a forward biased p–n junction diode and results in an offset voltage in the output characteristic. Current flow contributions are shown in Figure 3a using varying line thickness with the thicker lines indicating a high current path. For a fast device, the N+ buffer layer is highly doped for recombination and speedy turn off. The additional doping keeps the gain of the PNP low and allows two–thirds of the current to flow through the base of the PNP (electron current) while one–third passes through the collector (hole current). Rshorting is the parasitic resistance of the P+ emitter region. Current flowing through Rshorting can result in a voltage across the base–emitter junction of the NPN. If the base–emitter voltage is above a certain threshold level, the NPN will begin to conduct causing the NPN and PNP to enhance each other’s current flow and both devices can become saturated. This results in the device latching in a fashion similar to an SCR. Device processing directs currents within the device and keeps the voltage across Rshorting low to avoid latching. The IGBT can be gated off unlike the SCR which has to wait for the current to cease allowing recombination to take place in order to turn off. IGBTs offer an advantage over the SCR by controlling the current with the device, not the device with the current. The internal MOSFET of the IGBT when gated off will stop current flow and at that point, the stored charges can only be dissipated through recombination. The IGBT’s on–voltage is represented by sum of the offset voltage of the collector to base junction of the PNP transistor, the voltage drop across the modulated resistance Rmod and the channel resistance of the internal MOSFET. Unlike the MOSFET where increased temperature results in increased RDS(on) and increased forward voltage drop, the forward drop of an IGBT stays relatively unchanged at increased temperatures. Switching Speed Until recently, the feature that limited the IGBT from serving a wide variety of applications was its relatively slow turn–off speed when compared to a power MOSFET. While turn–on is fairly rapid, initial IGBTs had current fall times of around three microseconds. The turn–off time of an IGBT is slow because many minority carriers are stored in the n– epi region. When the gate is initially brought below the threshold voltage, the n– epi contains a very large concentration of electrons and there will be significant injection into the P+ substrate and a corresponding hole injection into the n– epi. As the electron concentration in the n–region decreases, the electron injection decreases, leaving the rest of the electrons to recombine. Therefore, the turn–off of an IGBT has two phases: an injection phase where the collector current falls very quickly, and a recombination phase in which the collector current decrease more slowly. Figure 5 shows the switching waveform and the tail time contributing factors of a “fast” IGBT designed for PWM motor control service. ÉÉÉÉ ÉÉÉÉ ÉÉÉÉ ÉÉÉÉ ÉÉÉÉ ÇÇÇÇ ÇÇ ÇÇ ÇÇ ÇÇ Figure 5. IGBT Current Turn–off Waveform 6 5 4 3 2 1 0 –1 0 200 400 600 800 1000 I C (AMPS) TAIL TIME of MOTOROLA GEN. 2 IGBT #2 in 1.0 hp MOTOR DRIVE at 1750 RPM PNP TURN–OFF PORTION TAIL TIME MOSFET TURN–OFF PORTION In power MOSFETs, the switching speed can be greatly affected by the impedance in the gate drive circuit. Efforts to minimize gate drive impedance for IGBTs are also recommended. Also, choose an optimal device based on switching speed or use a slower device with lower forward drop and employ external circuitry to enhance turn off. A turn–off mechanism is suggested in a paper by Baliga et al [2].
AN1541 A FINAL COMPARISON OF IGBTS BJTs AND Short Circuit Rated Devices POWER MOSFETS Using IGBTs in motor control environments requires the device to withstand short circuit current for a given period The conduction losses of BJTs and IGBTs is related to the Although this period varies with the application, a typical forward voltage drop of the device while MOSFETs determine value of ten microseconds is used for designing these conduction loss based on RDS(on). To get a relative comparison of turn-off time and conduction associated specialized IGBTs. Notice that this is only a typical value and it is suggested that the reader confirm the value given on the losses, data is presented in Table 1 where the on-resistances data sheet. IGBTs can be made to withstand short circuit of power MOSFET, IGBT and a BjT at junction temperatures of 25C and 150C are shown conditions by altering the device structure to include an Note that the devices in Table 1 have approximately the additional resistance(Re, in Figure 6)in the main current path The benefits associated with the additional series resistance same ratings. However, to achieve these ratings the chip size are twofold of the devices vary significantly. The bipolar transistor requires 1.2 times more silicon area than the igbt and the mosfet requires 2.2 times the area of the IGBT to achieve the same GATE ratings. This differences in die area directly impacts the cost EMITER f the product. At higher currents and at elevated temperatures, the IGBT offers low forward drop and a POLYSILICON GATE switching time similar to the BjT without the drive difficulties Table 1 confirms the findings offered earlier in Figure 1a and elaborates further to include a BJT comparison and temperature effects. The reduced power conduction losses offered by the IGBT lower power dissipation and heat sink Thermal Resistance An IGBT and power MOSFET produced from the same size die have similar junction-to-case thermal resistance because of their similar structures. The thermal resistance of a pow P+ SUBSTRATE MOSFET can be determined by testing for variations in temperature sensitive parameters(TSPs). These parameters are the source-to-drain diode on-voltage, the gate-to-source threshold voltage, and the drain-to-source on-resistance. All previous measurements of thermal resistance of power MOSFETs at Motorola were performe Figure 6. Cross Section and Equivalent Schemati using the source-to-drain diode as the TSP. Since an IGBT of a short Circuit Rated Insulated gate does not have an inverse parallel diode another TsP had to Bipolar Transistor Cell gate-to-emitter threshold voltage was used as the TSP to First, the voltage created across Re, by the large current measure the junction temperature of an IGBT to determine its passing through Re, increases the percentage of the gate thermal resistance. However before testing IGBTS, a voltage across Re, by the classic voltage divider equation correlation between the two test methods was established by Assuming the drive voltage applied to the gate-to-emitter comparing the test results of MOSFETs using both TSPs. By remains the same, the voltage actually applied across the testing for variations in threshold voltage, it was determined gate-to-source portion of the device is now lower, and the that the thermal resistance of MOSFETS and IGBTs are device is operating in an area of the transconductance curve essentially the same for devices with equivalent die size that reduces the gain and it will pass less current. Table 1. Advantages Offered by the IGBT When Comparing the MOSFET, IGBT and Bipolar Transistor On-Resistances (Over Junction Temperature )and Fall Times(Resistance Values at 10 Amps of Current) Characteristic TMOS Current Rating 20A 20A 20A Voltage Rating 500v 600V R(on)@ TJ=25C 0.29 0.24 0.18g Ron)@TJ=150°c 0.6g Fall Time(Typical 40 ns 200ns Indicates VcEo Rating MOTOROLA
MOTOROLA 5 A FINAL COMPARISON OF IGBTs, BJTs AND POWER MOSFETs The conduction losses of BJTs and IGBTs is related to the forward voltage drop of the device while MOSFETs determine conduction loss based on RDS(on). To get a relative comparison of turn–off time and conduction associated losses, data is presented in Table 1 where the on–resistances of a power MOSFET, an IGBT and a BJT at junction temperatures of 25°C and 150°C are shown. Note that the devices in Table 1 have approximately the same ratings. However, to achieve these ratings the chip size of the devices vary significantly. The bipolar transistor requires 1.2 times more silicon area than the IGBT and the MOSFET requires 2.2 times the area of the IGBT to achieve the same ratings. This differences in die area directly impacts the cost of the product. At higher currents and at elevated temperatures, the IGBT offers low forward drop and a switching time similar to the BJT without the drive difficulties. Table 1 confirms the findings offered earlier in Figure 1a and elaborates further to include a BJT comparison and temperature effects. The reduced power conduction losses offered by the IGBT lower power dissipation and heat sink size. Thermal Resistance An IGBT and power MOSFET produced from the same size die have similar junction–to–case thermal resistance because of their similar structures. The thermal resistance of a power MOSFET can be determined by testing for variations in temperature sensitive parameters (TSPs). These parameters are the source–to–drain diode on–voltage, the gate–to–source threshold voltage, and the drain–to–source on–resistance. All previous measurements of thermal resistance of power MOSFETs at Motorola were performed using the source–to–drain diode as the TSP. Since an IGBT does not have an inverse parallel diode, another TSP had to be used to determine the thermal resistance. The gate–to–emitter threshold voltage was used as the TSP to measure the junction temperature of an IGBT to determine its thermal resistance. However before testing IGBTs, a correlation between the two test methods was established by comparing the test results of MOSFETs using both TSPs. By testing for variations in threshold voltage, it was determined that the thermal resistance of MOSFETs and IGBTs are essentially the same for devices with equivalent die size . Short Circuit Rated Devices Using IGBTs in motor control environments requires the device to withstand short circuit current for a given period. Although this period varies with the application, a typical value of ten microseconds is used for designing these specialized IGBT’s. Notice that this is only a typical value and it is suggested that the reader confirm the value given on the data sheet. IGBTs can be made to withstand short circuit conditions by altering the device structure to include an additional resistance (Re, in Figure 6) in the main current path. The benefits associated with the additional series resistance are twofold. Figure 6. Cross Section and Equivalent Schematic of a Short Circuit Rated Insulated Gate Bipolar Transistor Cell POLYSILICON GATE N+ P+ N– EPI N+ BUFFER P+ SUBSTRATE P– Rmod EMITTER N+ P+ P– COLLECTOR GATE NPN MOSFET PNP KEY METAL SiO2 Rshorting Re First, the voltage created across Re, by the large current passing through Re, increases the percentage of the gate voltage across Re, by the classic voltage divider equation. Assuming the drive voltage applied to the gate–to–emitter remains the same, the voltage actually applied across the gate–to–source portion of the device is now lower, and the device is operating in an area of the transconductance curve that reduces the gain and it will pass less current. Table 1. Advantages Offered by the IGBT When Comparing the MOSFET, IGBT and Bipolar Transistor On–Resistances (Over Junction Temperature) and Fall Times (Resistance Values at 10 Amps of Current) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Characteristic ÁÁÁÁÁÁÁÁ TMOS ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ IGBT ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ Bipolar ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Current Rating ÁÁÁÁÁÁÁÁ 20 A ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 20 A ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 20 A ÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Voltage Rating Á ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ 500 V Á ÁÁÁÁÁÁÁ ÁÁÁÁÁ 600 V Á ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ 500 V* Á ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ R(on) @ TJ = 25°C ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ 0.2 Ω ÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 0.24 Ω ÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ 0.18 Ω ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ R(on) @ TJ = 150°C ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ 0.6 Ω ÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 0.23 Ω ÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 0.24 Ω** ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Fall Time (Typical) ÁÁÁÁÁÁÁÁ 40 ns ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 200 ns ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 200 ns * Indicates VCEO Rating ** BJT TJ = 100°C
AN1541 Second. the leveloped across Re results in a similar division ross Shorting and VBE of the NPN transistor ill be less likely to attain a VBe high enough to device on and cause a latch-up situation The two situations described work together to protect the device from catastrophic failure. The protection period is specified with the device ratings, allowing circuit designers VOLTAGE RM(rec) the time needed to detect a fault and shut off the device The introduction of the series resistance Re also results in additional power loss in the device by slightly elevating the forward drop of the device. However, the magnitude of short circuit current is large enough to require a very low Re value The additional conduction loss of the device due to the presence of Re is not excessive when comparing a short circuit rated iGBT to a non-short circuit rated device Anti-Parallel Diode When using IGBTs for motor control, designers have to place a diode in anti-parallel across the device in order to handle the regenerative or inductive currents of the motor. As Figure 7. Waveforms Associated with discussed earlier. due to structural differences the igbt does Anti-Parallel Diode Turn-off not have a parasitic diode like that found in a MOSFET. Designers found that the diode within the MOSFET was, in fact, a parasitic, i.e., not optimized i n the design process, and its performance was poor for use as a current recovery device APPLICATION OF IGBTs: due to slow switching speed. To overcome the lack of PULSE WIDTH MODULATED INDUCTION performance, an optimized anti-parallel diode was used MOTOR DRIVE APPLICATION across the MOSFET source-to-drain. Placing a packaged diode external to the MOSFET itself created performance operated, pulse-width modul problems due to the switching delays resulting from the motor drives are an application well suited for IGBTs. In this parasitics introduced by the packages. The optimal setup is to application, as shown in Figure 8, IGBTs are used as the have the diode copackaged with the device. A specific line of power switch to PWM the voltage supplied to a motor to control its speed GBTs has been created by Motorola to address this issue These devices work very well in applications where energy is Depending on the application, the igBT may be required to recovered to the source and are favored by motor control operate from a full-wave rectified line. This can require devices to have six hundred volt ratings for 230 VAC line voltage inputs, and twelve hundred volt ratings for 575 VAC Like the switching device itself, the anti-parallel diode volt line inputs. IGBTs that block high voltage offer fast should exhibit low leakage current, low forward voltage dro and fast switching speed. As shown in Figure 7, the diode switching and low conduction losses, and allow for the design forward drop multiplied by the average current it passes is the of efficient, high frequency drives of this type Devices used in total conduction loss produced. In addition, large reverse motor drive applications must be robust and capable of recovery currents can escalate switching losses. A detailed withstanding faults long enough for a protection scheme to be activated Short circuit rated devices offer safe reliable motor planation of reverse Appendix. A secondary effect caused by large reverse drive operation recovery currents is generated EMI at both the switching CONCLUSION frequency and the frequency of the resulting ringing aveform. This EMI requires additional filtering to be The IGBT is a one of several options for designers to choose designed into the circuit By copackaging parts, the parasitic from for power control in switching applications. The features inductances that contribute to the ringing are greatly reduced Also, copackaged products can be used in designs to reduce ease of drive and relatively fast switching speeds makes it a power dissipation and increase design efficiency technology of choice for moderate speed, high voltage applications. New generations of devices will reduce the on-resistance, increase speed and include levels of integration that simplify protection schemes and device drive requirements. The reliability and performance advantages of IGB Ts are value added traits that offer circuit designers energy efficient options at reduced costs MOTOROLA
6 MOTOROLA Second, the voltage developed across Re results in a similar division of voltage across Rshorting and VBE of the NPN transistor. The NPN will be less likely to attain a VBE high enough to turn the device on and cause a latch–up situation. The two situations described work together to protect the device from catastrophic failure. The protection period is specified with the device ratings, allowing circuit designers the time needed to detect a fault and shut off the device. The introduction of the series resistance Re also results in additional power loss in the device by slightly elevating the forward drop of the device. However, the magnitude of short circuit current is large enough to require a very low Re value. The additional conduction loss of the device due to the presence of Re is not excessive when comparing a short circuit rated IGBT to a non–short circuit rated device. Anti–Parallel Diode When using IGBT’s for motor control, designers have to place a diode in anti–parallel across the device in order to handle the regenerative or inductive currents of the motor. As discussed earlier, due to structural differences the IGBT does not have a parasitic diode like that found in a MOSFET. Designers found that the diode within the MOSFET was, in fact, a parasitic, i.e., not optimized in the design process, and its performance was poor for use as a current recovery device due to slow switching speed. To overcome the lack of performance, an optimized anti–parallel diode was used across the MOSFET source–to–drain. Placing a packaged diode external to the MOSFET itself created performance problems due to the switching delays resulting from the parasitics introduced by the packages. The optimal setup is to have the diode copackaged with the device. A specific line of IGBTs has been created by Motorola to address this issue. These devices work very well in applications where energy is recovered to the source and are favored by motor control designers. Like the switching device itself, the anti–parallel diode should exhibit low leakage current, low forward voltage drop and fast switching speed. As shown in Figure 7, the diode forward drop multiplied by the average current it passes is the total conduction loss produced. In addition, large reverse recovery currents can escalate switching losses. A detailed explanation of reverse recovery can be found in the Appendix. A secondary effect caused by large reverse recovery currents is generated EMI at both the switching frequency and the frequency of the resulting ringing waveform. This EMI requires additional filtering to be designed into the circuit. By copackaging parts, the parasitic inductances that contribute to the ringing are greatly reduced. Also, copackaged products can be used in designs to reduce power dissipation and increase design efficiency. Figure 7. Waveforms Associated with Anti–Parallel Diode Turn–off IRM(rec) TIME TIME TIME POWER VOLTAGE CURRENT IIGBT IDIODE Vf APPLICATION OF IGBTs: PULSE WIDTH MODULATED INDUCTION MOTOR DRIVE APPLICATION Line–operated, pulse–width modulated, variable–speed motor drives are an application well suited for IGBTs. In this application, as shown in Figure 8, IGBTs are used as the power switch to PWM the voltage supplied to a motor to control its speed. Depending on the application, the IGBT may be required to operate from a full–wave rectified line. This can require devices to have six hundred volt ratings for 230 VAC line voltage inputs, and twelve hundred volt ratings for 575 VAC volt line inputs. IGBTs that block high voltage offer fast switching and low conduction losses, and allow for the design of efficient, high frequency drives of this type. Devices used in motor drive applications must be robust and capable of withstanding faults long enough for a protection scheme to be activated. Short circuit rated devices offer safe, reliable motor drive operation. CONCLUSION The IGBT is a one of several options for designers to choose from for power control in switching applications. The features of the IGBT such as high voltage capability, low on–resistance, ease of drive and relatively fast switching speeds makes it a technology of choice for moderate speed, high voltage applications. New generations of devices will reduce the on–resistance, increase speed and include levels of integration that simplify protection schemes and device drive requirements. The reliability and performance advantages of IGBTs are value added traits that offer circuit designers energy efficient options at reduced costs.
AN1541 NDUCTIO MOTOR TEMPERATURE MIXED MODE IC CUSTOM LINEAR CONTROL OR ASIC SYSTEM STANDARD CELL OPTO& LVIC PHASE CURRENTS AND VOLTAGES Figure 8. Typical Pulse-Width, Modulated, Variable-Speed Induction Motor Drives Are Where IGBTs Offer Performance Advantages ACKNOWLEDGEMENTS REFERENCES The writing of this document was assisted by a number of [1] D. Y. chen, J. Yang, and J. Lee "Application of the ternal device designers. Their assistance was greatly IGT/COMFET to Zero-Current appreciated by the authors. Bill Fragale, Steve Robb and Converters, "PESC, 1987 Vasudev Venkatesan provided device operation insight and [2 ]B J Baliga, "Analysis of Insulated Gate Transistor Turn-off reference materials. Graphic material was provided by Basal Characteristics, IEEE Electron Device Lett. EDL-6, (1985) Almesfer and Steve Robb. Finally, C S Mitter assisted with pp 74-77 editing and accuracy of the material [3]BJ Baliga, "Switching Speed Enhancement in Insulated Gate Transistors by Electron Irradiation, "IEEE Transactions on Electron Devices, ED-31, (1984), pp 1790-1795 MOTOROLA
MOTOROLA 7 IGBT 1/2 BRIDGE IGBT 1/2 BRIDGE Figure 8. Typical Pulse–Width, Modulated, Variable–Speed Induction Motor Drives Are Where IGBT’s Offer Performance Advantages 230 VAC TEMPERATURE CONTROL SYSTEM I/O LON CONTROL IC MCU OR ASIC MIXED MODE IC CUSTOM LINEAR OR STANDARD CELL GATE DRIVE HVIC OR OPTO & LVIC DIODE BRIDGE FILTER CAPACITOR INDUCTION MOTOR IGBT 1/2 BRIDGE PHASE CURRENTS AND VOLTAGES ACKNOWLEDGEMENTS The writing of this document was assisted by a number of internal device designers. Their assistance was greatly appreciated by the authors. Bill Fragale, Steve Robb and Vasudev Venkatesan provided device operation insight and reference materials. Graphic material was provided by Basam Almesfer and Steve Robb. Finally, C. S. Mitter assisted with editing and accuracy of the material. REFERENCES [1] D. Y. Chen, J. Yang, and J. Lee “Application of the IGT/COMFET to Zero–Current Switching Resonant Converters,” PESC, 1987. [2] B. J. Baliga, “Analysis of Insulated Gate Transistor Turn–off Characteristics,” IEEE Electron Device Lett. EDL–6, (1985), pp. 74–77 . [3] B. J. Baliga, “Switching Speed Enhancement in Insulated Gate Transistors by Electron Irradiation,” IEEE Transactions on Electron Devices, ED–31, (1984), pp. 1790–1795.
AN APPENDIX Diode Reverse Recovery Analysis [4] a typical reverse recovery waveform is shown in Figure A-1. The reverse recovery time trr has been traditionally defined as the time from diode current zero-crossing to where the current returns to within 10% of the peak recovery current IRM(rec). This does not give enough information to fully characterize the waveform shape abetter way to characterize the rectifier reverse recovery is to partition the reverse recovery time into two different regions ta and tb as shown in Figure A-1. The ta time is a function of the forward current and the applied di/dt. a charge can be assigned to this region denoted Qa, the area under the curve. The tb portion of the reverse recovery current is not very well understood Measured tb times vary greatly with the switch characteristic, circuit parasitics, load inductance and the applied reverse voltage. a relative softness can be defined as the ratio of tb to ta. General purpose rectifiers are very soft(softness factor of about 1.0), fast recovery diodes are fairly soft(softness factor of about 0.5)and ultrafast rectifiers are very abrupt (softness factor of about 0.2) tr total reverse recovery time [4] Source Controls. TMOs Power MOSFET ta fall time due to stored minority charge Transistor Data, Q4/92, DL135, Rev 4,(Phoenix: Motorola t= application and device dependent nc,1992),pp.2-922to2-9-23 RM(rec)=peak reverse recovery current Figure A-1. Reverse Recovery Waveform up AUEA al parameters can and do var ling"Typicals" must be validated for each customer application by customers not convey any license under its patent rights nor the rights of others. any other he Motorola product could create a situation w or use Motorola products for any suc tly or indirectly, any claim of personal injury or death such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and (AA) are registered trademarks of Motorola, Inc. Motorola, Inc is an Equal Opportunity/Affirmative Action Employer. SA/EUROPE: Motorola Literature Distribution N: Nippon Motorola Ltd. Tatsum-SPD-JLDC, Toshikatsu O P.O. Box 20912: Phoenix, Arizona 85036. 1-800-441-2447 bu-Butsuryu-Center, 3-14-2 Tatsumi Koto-Ku, Tokyo 135, Japan. 03-3521-8315 MFAX:RMFAXO@email.sps.mot.com-TOUCHTONE(602)244-6609HONGKONG:MotorolaSemiconductorsHKLtd.8BTaiPingIndustrialPark, 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852-26629298 MTO-OLA AN1541
8 MOTOROLA APPENDIX Diode Reverse Recovery Analysis [4] Figure A–1. Reverse Recovery Waveform total reverse recovery time fall time due to stored minority charge application and device dependent peak reverse recovery current ÉÉÉ ÉÉÉ ÉÉÉ ÉÉÉ ÉÉÉ tb ta trr IRM(rec) IF di/dt Qa Qb trr = ta = tb = IRM(rec) = A typical reverse recovery waveform is shown in Figure A–1. The reverse recovery time trr has been traditionally defined as the time from diode current zero–crossing to where the current returns to within 10% of the peak recovery current IRM(rec). This does not give enough information to fully characterize the waveform shape. A better way to characterize the rectifier reverse recovery is to partition the reverse recovery time into two different regions, ta and tb, as shown in Figure A–1. The ta time is a function of the forward current and the applied di/dt. A charge can be assigned to this region denoted Qa, the area under the curve. The tb portion of the reverse recovery current is not very well understood. Measured tb times vary greatly with the switch characteristic, circuit parasitics, load inductance and the applied reverse voltage. A relative softness can be defined as the ratio of tb to ta. General purpose rectifiers are very soft (softness factor of about 1.0), fast recovery diodes are fairly soft (softness factor of about 0.5) and ultrafast rectifiers are very abrupt (softness factor of about 0.2). [4] Source: “Motor Controls,” TMOS Power MOSFET Transistor Data, Q4/92, DL135, Rev 4, (Phoenix: Motorola, Inc., 1992), pp. 2–9–22 to 2–9–23. Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters can and do vary in different applications. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer. How to reach us: USA / EUROPE: Motorola Literature Distribution; JAPAN: Nippon Motorola Ltd.; Tatsumi–SPD–JLDC, Toshikatsu Otsuki, P.O. Box 20912; Phoenix, Arizona 85036. 1–800–441–2447 6F Seibu–Butsuryu–Center, 3–14–2 Tatsumi Koto–Ku, Tokyo 135, Japan. 03–3521–8315 MFAX: RMFAX0@email.sps.mot.com – TOUCHTONE (602) 244–6609 HONG KONG: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park, INTERNET: http://Design–NET.com 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298 AN1541/D ◊