《电子工程师手册》学习资料（英文版）chapter 39 Microwave Devices.pdf_大学文库

39 Microwave Devices Characterization of passive elements Transmission Line Sections.Discontinuities. Impedance Transformers Terminations. Attenuators. Microwave Resonators. Tuning Michael B. steer ements. Hybrid Circuits and Directional Couplers. Filters Ferrite Components.Passive Semiconductor Devices 39.2 Active microwave Devices Robert ].Trew Semiconductor Material Properties. Two-Terminal Active Case Western Reserve University Microwave Devices Three-Terminal Active Microwave Devices 39.1 Passive microwave devices Michael B Steer Wavelengths in air at microwave and millimeter-wave frequencies range from 1 m at 300 MHz to 1 mm at 300 GHz and are comparable to the physical dimensions of fabricated electrical components. For this reason circuit components commonly used at lower frequencies, such as resistors, capacitors, and inductors, are not readily available above 10 GHz. The available microwave frequency lumped elements have dimensions of around mm. The relationship between the wavelength and physical dimensions enables new classes of distributed components to be constructed that have no analogy at lower frequencies. Components are realized by disturbing the field structure on a transmission line, resulting in energy storage and thus reactive effects. Electric(E)field isturbances have a capacitive effect and the magnetic(H)field disturbances appear inductive. Microwave components are fabricated in waveguide, coaxial lines, and strip lines. The majority of circuits are constructed sing strip lines as the cost is relatively low and they are highly reproducible due to the photolithographic hniques used. Fabrication of waveguide components requires precision machining but they can tolerate higher power levels and are more easily realized at millimeter-wave frequencies(30-300 GHz)than either coaxial or microstrip components. Characterization of Passive elements Passive microwave elements are defined in terms of their reflection and transmission properties for an incident wave of electric field or voltage Scattering(S)parameters are based on traveling waves and so naturally describe hese properties. As well they are the only ones that can be measured directly at microwave frequencies. s parameters are defined in terms of root power waves which in turn are defined using forward and backward traveling voltage waves. Consider the N port network of Fig 39.1 where the nth port has a reference transmission ine of characteristic impedance Zo and of infinitesimal length. The transmission line at the nth port serves to separate the forward and backward traveling voltage(V+ and Vm)and current(It and In)waves. The reference characteristic impedance matrix, Z, is a diagonal matrix, Z,= diag(Zol-.Zom. ZoN), and the root power waves at the nth port, a, and bn, are defined by c 2000 by CRC Press LLC

© 2000 by CRC Press LLC 39 Microwave Devices 39.1 Passive Microwave Devices Characterization of Passive Elements • Transmission Line Sections • Discontinuities • Impedance Transformers • Terminations • Attenuators • Microwave Resonators • Tuning Elements • Hybrid Circuits and Directional Couplers • Filters • Ferrite Components • Passive Semiconductor Devices 39.2 Active Microwave Devices Semiconductor Material Properties • Two-Terminal Active Microwave Devices • Three-Terminal Active Microwave Devices 39.1 Passive Microwave Devices Michael B. Steer Wavelengths in air at microwave and millimeter-wave frequencies range from 1 m at 300 MHz to 1 mm at 300 GHz and are comparable to the physical dimensions of fabricated electrical components. For this reason circuit components commonly used at lower frequencies, such as resistors, capacitors, and inductors, are not readily available above 10 GHz. The available microwave frequency lumped elements have dimensions of around 1 mm. The relationship between the wavelength and physical dimensions enables new classes of distributed components to be constructed that have no analogy at lower frequencies. Components are realized by disturbing the field structure on a transmission line, resulting in energy storage and thus reactive effects. Electric (E) field disturbances have a capacitive effect and the magnetic (H) field disturbances appear inductive. Microwave components are fabricated in waveguide, coaxial lines, and strip lines. The majority of circuits are constructed using strip lines as the cost is relatively low and they are highly reproducible due to the photolithographic techniques used. Fabrication of waveguide components requires precision machining but they can tolerate higher power levels and are more easily realized at millimeter-wave frequencies (30–300 GHz) than either coaxial or microstrip components. Characterization of Passive Elements Passive microwave elements are defined in terms of their reflection and transmission properties for an incident wave of electric field or voltage. Scattering (S) parameters are based on traveling waves and so naturally describe these properties. As well they are the only ones that can be measured directly at microwave frequencies. S parameters are defined in terms of root power waves which in turn are defined using forward and backward traveling voltage waves. Consider the N port network of Fig. 39.1 where the nth port has a reference transmission line of characteristic impedance Z0n and of infinitesimal length. The transmission line at the nth port serves to separate the forward and backward traveling voltage (Vn + and Vn – ) and current (In + and I n –) waves. The reference characteristic impedance matrix, Z0 is a diagonal matrix, Z0 = diag(Z01…Z0n…Z0N), and the root power waves at the nth port, an and bn , are defined by a V Z b V Z (39.1) n n n n n n = = + 0 0 and – Michael B. Steer North Carolina State University Robert J. Trew Case Western Reserve University

© 2000 by CRC Press LLC The reflection characteristics are also described by the voltage standing wave ratio (VSWR), a quantity that can be measured using relatively simple equipment. The VSWR is the ratio of the maximum voltage amplitude on the imput transmission line to the minimum voltage amplitude . Thus, TABLE 39.1 Two-Port S Parameter Conversion Chart for Impedance, Z, Admittance, Y, and Hybrid, H, Parameters S In Terms of S Z Y H Note: The Z¢, Y¢ and H¢ parameters are normalized to Z0. FIGURE 39.2 Incident, reflected and transmitted traveling voltage waves at (a) a passive microwave element and (b) a transmission line. z zZ z zZ ¢ = ¢ = 11 11 0 12 12 0 z zZ z zZ ¢ = ¢ = 21 21 0 22 22 0 d = ( ) Z Z ZZ ¢ + ( ) ¢ + - ¢ ¢ 11 22 12 21 1 1 d= - ( ) 1 1( ) - - 11 22 12 21 S S SS S Z Z ZZ 11 11 22 12 21 = ( ) ¢ - 1 1 ( ) ¢ + - ¢ ¢ [ ] d ¢ = + ( )( ) - + [ ] z S S SS 11 11 22 12 21 1 1 d S Z 12 12 = 2 ¢ d Z S ¢ = 12 12 2 d S Z 21 21 = 2 ¢ d Z S ¢ = 21 21 2 d S Z Z ZZ 22 11 22 12 21 = ( ) ¢ + 1 1 ( ) ¢ - - ¢ ¢ [ ] d ¢ = - ( )( ) + + Z S S SS 22 11 22 12 21 [ ] 1 1 d Y YZ Y YZ ¢ = ¢ = 11 11 0 12 12 0 Y YZ Y YZ ¢ = ¢ ¢ = 21 21 0 22 22 0 d= + ( ) 1 1 ¢ ( ) + ¢ = ¢ ¢ 11 22 12 21 Y Y YY d= + ( ) 1 1( ) + - 11 22 12 21 S S SS S Y Y YY 11 11 22 12 21 = - ( ) 1 1 ¢ ( ) + ¢ + ¢ ¢ [ ] d ¢ = - ( )( ) + + Y S S SS 11 11 22 12 21 [ ] 1 1 d S Y 12 12 = -2 ¢ d Y S ¢ = - 12 12 2 d S Y 21 21 = -2 ¢ d Y S ¢ = - 21 21 2 d S Y Y YY 22 11 22 12 21 = + ( ) 1 1 ¢ ( ) - ¢ + ¢ ¢ [ ] d ¢ = + ( )( ) - + Y S S SS 22 11 22 12 21 [ ] 1 1 d H HZ H H ¢ = ¢ = 11 11 0 12 12 H H H HZ ¢ = ¢ = 21 21 22 22 0 d= + ( ) 1 1 ¢ ( ) + ¢ - ¢ ¢ H H HH 11 22 12 21 d= - ( ) 1 1( ) + + 11 22 12 21 S S SS S H H HH 11 11 22 12 21 = ( ) ¢ - 1 1 ( ) ¢ + - ¢ ¢ [ ] d ¢ = + ( )( ) + - H S S SS 11 11 22 12 21 [ ] 1 1 d S H 12 12 = 2 ¢ d H S ¢ = 12 12 2 d S H 21 21 = -2 ¢ d H S ¢ = - 21 21 2 d S H H HH 22 11 22 12 21 = + ( ) 1 1 ¢ ( ) - ¢ + ¢ ¢ [ ] d ¢ = - ( )( ) - - H S S SS 22 11 22 12 21 [ ] 1 1 d V V 1 1 + - ( ) + V V 1 1 + - ( ) - VSWR = 1 1 1 1 ( + G G ) ( - )

SETFLCE DIELECTRIC FIGURE 39. 3 Sections of transmission lines used for interconnecting components:(a)waveguide tapered section (b)waveguide E-plane bend, (c)waveguide H-plane bend, (d)waveguide twist, and (e)microstrip taper. Most passive devices, with the notable exception of ferrite devices, are reciprocal and so Spy=Sap. A loss-less sp 1, which is a statement of power cons Most microwave circuits are designed to minimize the reflected energy and maximize transmission at least over the frequency range of operation. Thus, the return loss is high and the VSwR =1 for well-designed circuits. A terminated transmission line such as that in Fig. 39.2(b) has an input impedance z.=Z ZL+ jZo tanh yd Zo+ jz, tanh yo Thus, a short section(Ya < 1)of a short circuited(Z=0)transmission line looks like an inductor and capacitor if it is open circuited(Z,= oo). When the line is a half wavelength long, an open circuit is at the input to the line if the other end is short circuited Transmission line sections The simplest microwave circuit element is a uniform section of transmission line which can be used to introduce a time delay or a frequency-dependent phase shift. Other line segments for interconnections include bend corners,twists, and transitions between lines of different dimensions(see Fig 39.3). The dimensions and shapes are designed to minimize reflections and so maximize return loss and minimize insertion loss Discontinuities The waveguide discontinuities shown in Fig. 39. 4(a)-(f) illustrate most clearly the use of E and H field distu bances to realize capacitive and inductive components. An E-plane discontinuity[ Fig. 39. 4(a)] can be modeled approximately by a frequency-dependent capacitor. H-plane discontinuities [Figs. 39.4(b)and (c)] resemble inductors as does the circular iris of Fig. 39.4(d). The resonant waveguide iris of Fig. 39.4(e) disturbs both the E and H fields and can be modeled by a parallel LC resonant circuit near the frequency of resonance Posts in waveguide are used both as reactive elements [Fig. 39.4(f)] and to mount active devices [Fig. 39.4(g).The equivalent circuits of microstrip discontinuities [Figs. 39.4(k)-o)) are again modeled by capacitive elements if the E field is interrupted and by inductive elements if the H field(or current)is interrupted. The stub shown in Fig. 39.4() presents a short circuit to the through transmission line when the length of the stub is / 4.When the stubs are electrically short(<< A 4)they introduce shunt capacitances in the through transmission line

© 2000 by CRC Press LLC Most passive devices, with the notable exception of ferrite devices, are reciprocal and so Spq = Sqp. A loss-less passive device also satisfies the unitary condition: which is a statement of power conservation indicating that all power is either reflected or transmitted. Most microwave circuits are designed to minimize the reflected energy and maximize transmission at least over the frequency range of operation. Thus, the return loss is high and the VSWR ª 1 for well-designed circuits. A terminated transmission line such as that in Fig. 39.2(b) has an input impedance Thus, a short section (gd << 1) of a short circuited (ZL = 0) transmission line looks like an inductor and a capacitor if it is open circuited (ZL = •). When the line is a half wavelength long, an open circuit is presented at the input to the line if the other end is short circuited. Transmission Line Sections The simplest microwave circuit element is a uniform section of transmission line which can be used to introduce a time delay or a frequency-dependent phase shift. Other line segments for interconnections include bends, corners, twists, and transitions between lines of different dimensions (see Fig. 39.3). The dimensions and shapes are designed to minimize reflections and so maximize return loss and minimize insertion loss. Discontinuities The waveguide discontinuities shown in Fig. 39.4(a)–(f) illustrate most clearly the use of E and H field disturbances to realize capacitive and inductive components. An E-plane discontinuity [Fig. 39.4(a)] can be modeled approximately by a frequency-dependent capacitor. H-plane discontinuities [Figs. 39.4(b) and (c)] resemble inductors as does the circular iris of Fig. 39.4(d). The resonant waveguide iris of Fig. 39.4(e) disturbs both the E and H fields and can be modeled by a parallel LC resonant circuit near the frequency of resonance. Posts in waveguide are used both as reactive elements [Fig. 39.4(f)] and to mount active devices [Fig. 39.4(g)]. The equivalent circuits of microstrip discontinuities [Figs. 39.4(k)–(o)] are again modeled by capacitive elements if the E field is interrupted and by inductive elements if the H field (or current) is interrupted. The stub shown in Fig. 39.4(j) presents a short circuit to the through transmission line when the length of the stub is lg/4. When the stubs are electrically short (<< lg/4) they introduce shunt capacitances in the through transmission line. FIGURE 39.3 Sections of transmission lines used for interconnecting components: (a) waveguide tapered section, (b) waveguide E-plane bend, (c) waveguide H-plane bend, (d) waveguide twist, and (e) microstrip taper. Sp pq S 2 = 1, Z Z Z jZ d Z jZ d L L in = + + 0 0 0 tanh tanh g g

SVERSE H-PLAN HI+ 山 FIGURE 39.4 Discontinuities Waveguide discontinuities:(a)capacitive E-plane discontinuity,(b)inductive H-plane dis- continuity,(c)symmetrical inductive H-plane discontinuity,(d)inductive post discontinuity,(e)resonant window discon- tinuity,(f)capacitive post discontinuity(g) diode post mount, and(h)quarter-wave impedance transformer. Microstrip discontinuities:(i)quarter-wave impedance transformer, ()open microstrip stub, (k)step, (I)notch, (m)gap, (n)crossover, and(o)bend. Impedance Transformers Impedance transformers are used to interface two sections of line with different characteristic impedances The smoothest transition and the one with the broadest bandwidth is a tapered line as shown in Fig. 39. 3(a) and(e). This element tends to be very long and so step terminations called quarter-wave impedance transformers ee Fig. 39.4(h)and (i) are sometimes used although their bandwidth is relatively small centered on the frequency at which I=n 4. Ideally, Zo Terminations In a termination, power is absorbed by a length of lossy material at the end of a shorted piece of transmission line [Fig. 39.5(a)and(c)). This type of termination is called a matched load as power is absorbed and reflections are very small irrespective of the characteristic impedance of the transmission line. This is generally preferred the characteristic impedance of transmission lines varies with frequency, particularly so for waveguides. When the characteristic impedance of a line does not vary much with frequency, as is the case with a coaxia line, a simpler smaller termination can be realized by placing a resistor to ground [Fig. 39.5(b) Attenuators Attenuators reduce the level of a signal traveling along a transmission line. The basic construction is to make the line lossy but with a characteristic impedance approximating that of the connecting lines so as to reduce reflections. The line is made lossy by introducing a resistive vane in the case of a waveguide [Fig. 39.5(d)] replacing part of the outer conductor of a coaxial line by resistive material [Fig. 39.5(e)l, or covering the line into the transmission line is controlled, a variable attenuator is achieved, e.g, Fig. 39 material introduced by resistive material in the case of a microstrip line [Fig. 39.5(f). If the amount of lossy c 2000 by CRC Press LLC

© 2000 by CRC Press LLC Impedance Transformers Impedance transformers are used to interface two sections of line with different characteristic impedances. The smoothest transition and the one with the broadest bandwidth is a tapered line as shown in Fig. 39.3(a) and (e). This element tends to be very long and so step terminations called quarter-wave impedance transformers [see Fig. 39.4(h) and (i)] are sometimes used although their bandwidth is relatively small centered on the frequency at which l = lg/4. Ideally, Z0,2 = Terminations In a termination, power is absorbed by a length of lossy material at the end of a shorted piece of transmission line [Fig. 39.5 (a) and (c)]. This type of termination is called a matched load as power is absorbed and reflections are very small irrespective of the characteristic impedance of the transmission line. This is generally preferred as the characteristic impedance of transmission lines varies with frequency, particularly so for waveguides. When the characteristic impedance of a line does not vary much with frequency, as is the case with a coaxial line, a simpler smaller termination can be realized by placing a resistor to ground [Fig. 39.5(b)]. Attenuators Attenuators reduce the level of a signal traveling along a transmission line. The basic construction is to make the line lossy but with a characteristic impedance approximating that of the connecting lines so as to reduce reflections. The line is made lossy by introducing a resistive vane in the case of a waveguide [Fig. 39.5(d)], replacing part of the outer conductor of a coaxial line by resistive material [Fig. 39.5(e)], or covering the line by resistive material in the case of a microstrip line [Fig. 39.5(f)]. If the amount of lossy material introduced into the transmission line is controlled, a variable attenuator is achieved, e.g., Fig. 39.5(d). FIGURE 39.4 Discontinuities. Waveguide discontinuities: (a) capacitive E-plane discontinuity, (b) inductive H-plane discontinuity, (c) symmetrical inductive H-plane discontinuity, (d) inductive post discontinuity, (e) resonant window discontinuity, (f) capacitive post discontinuity, (g) diode post mount, and (h) quarter-wave impedance transformer. Microstrip discontinuities: (i) quarter-wave impedance transformer, (j) open microstrip stub, (k) step, (l) notch, (m) gap, (n) crossover, and (o) bend. Z Z 0, , 1 0 3

BANDWIDTH TER-WAVE CAVITY f R RADIATION TECTOR IGURE 39.6 Microwave resonators:(a)resonator response,(b)rectangular cavity resonator,(c)microstrip patch icrostrip gap-coupled reflection resonator,(e)transmission dielectric transmission resonator in microstrip, (f)parallel equivalent circuits, (g) series equivalent circuits, and (h)waveguide wavemeter HNG-LOADED PLUNGE TABLE FIGURE 39.7 Tuning elements:(a)waveguide sliding short circuit, ( b)coaxial line slug tuner, (c)microstrip stub with tuning pads This resonator has large coupling to the external circuit. The coupling can be reduced and photolithographically controlled by introducing a gap as shown in Fig. 39.6(d) for a microstrip gap-coupled transmission line reflection resonator. The Q of a resonator can be dramatically increased by using a high dielectric constant material as shown in Fig. 39.6(e) for a dielectric transmission resonator in microstrip. One simple application of a cavity resonator is the waveguide wavemeter[Fig. 39.6(h)). Here the resonant frequency of a rectangular cavity is varied by changing the physical dimensions of the cavity with a null of the detector indicating that the frequency corresponds to the resonant cavity frequency Tuning Elements In rectangular waveguide the basic adjustable tuning element is the sliding short shown in Fig. 39.7(a). Varying the position of the short will change resonance frequencies of cavities. It can be combined with hybrid tees to achieve a variety of tuning functions. The post in Fig. 39.4()can be replaced by a screw to obtain a screw tuner which is commonly used in waveguide filters. Sliding short circuits can be used in coaxial lines and in conjunction with branching elements to obtain stub tuners. Coaxial slug tuners are also used to provide adjustable matching at the input and output of active circuits. The slug is movable and changes the characteristic pedance of the transmission line. It is more difficult to achieve variable tuning in passive microstrip circuits One solution is to provide a number of pads as shown in Fig. 39.7(c) which, in this case, can be bonded to the stub to obtain an adjustable stub length. Variable amounts of phase shift can be inserted by using a variable length of line called a line stretcher, or by a line with a variable propagation constant. One type of waveguide variable phase shifter is similar to the variable attenuator of Fig. 39.5(d) with the resistive material replaced by a low-loss dielectric c 2000 by CRC Press LLC

© 2000 by CRC Press LLC This resonator has large coupling to the external circuit. The coupling can be reduced and photolithographically controlled by introducing a gap as shown in Fig. 39.6(d) for a microstrip gap-coupled transmission line reflection resonator. The Q of a resonator can be dramatically increased by using a high dielectric constant material as shown in Fig. 39.6(e) for a dielectric transmission resonator in microstrip. One simple application of a cavity resonator is the waveguide wavemeter [Fig. 39.6(h)]. Here the resonant frequency of a rectangular cavity is varied by changing the physical dimensions of the cavity with a null of the detector indicating that the frequency corresponds to the resonant cavity frequency. Tuning Elements In rectangular waveguide the basic adjustable tuning element is the sliding short shown in Fig. 39.7(a). Varying the position of the short will change resonance frequencies of cavities. It can be combined with hybrid tees to achieve a variety of tuning functions. The post in Fig. 39.4(f) can be replaced by a screw to obtain a screw tuner which is commonly used in waveguide filters. Sliding short circuits can be used in coaxial lines and in conjunction with branching elements to obtain stub tuners. Coaxial slug tuners are also used to provide adjustable matching at the input and output of active circuits. The slug is movable and changes the characteristic impedance of the transmission line. It is more difficult to achieve variable tuning in passive microstrip circuits. One solution is to provide a number of pads as shown in Fig. 39.7(c) which, in this case, can be bonded to the stub to obtain an adjustable stub length. Variable amounts of phase shift can be inserted by using a variable length of line called a line stretcher, or by a line with a variable propagation constant. One type of waveguide variable phase shifter is similar to the variable attenuator of Fig. 39.5(d) with the resistive material replaced by a low-loss dielectric. FIGURE 39.6 Microwave resonators: (a) resonator response, (b) rectangular cavity resonator, (c) microstrip patch resonator, (d) microstrip gap-coupled reflection resonator, (e) transmission dielectric transmission resonator in microstrip, (f) parallel equivalent circuits, (g) series equivalent circuits, and (h) waveguide wavemeter. FIGURE 39.7 Tuning elements: (a) waveguide sliding short circuit, (b) coaxial line slug tuner, (c) microstrip stub with tuning pads