复旦大学：微电子工程教学资源（参考论文）A 900 MHz, 21 dBm CMOS linear power amplifier with 35% PAE for RFID readers

Vol.31,No.12 Journal of Semiconductors December 2010 A 900 MHz,21 dBm CMOS linear power amplifier with 35%PAE for RFID readers* Han Kefeng(韩科锋)，Cao Shengguo(曹圣国)，Tan Xi(谈熙)'，Yan Na(闫娜)，Wang Junyu(王俊宇)， Tang Zhangwen(唐长文)，and Min Hao(闵昊) (State Key Laboratory of ASIC System,Fudan University.Shanghai 201203,China) Abstract:A two-stage differential linear power amplifier(PA)fabricated by 0.18 um CMOS technology is presented. An output matching and harmonic termination network is exploited to enhance the output power,efficiency and har- monic performance.Measurements show that the designed PA reaches a saturated power of 21.1 dBm and the peak power added efficiency(PAE)is 35.4%,the power gain is 23.3 dB from a power supply of 1.8 Vand the harmonics are well controlled.The total area with ESD protected PAD is 1.2 x 0.55 mm2.System measurements also show that this power amplifier meets the design specifications and can be applied for RFID reader. Key words:CMOS;power amplifier;PAE;power matching;RFID;reader D0:10.1088/1674-4926/31/12/125005 EEACC:1220 ity problem,a cascode topology is employed.To gain a power 1.Introduction of more than 20 dBm from a supply of 1.8 V,output matching for this differential PA is analyzed in detail and realized on a During the past several decades,CMOS has achieved great PCB board.To gain a higher efficiency and maintain relatively success for low cost and high yield,and it is also a popular good linearity,the PA works in deep class-AB mode.Measure- choice of system on chip for radio frequency (RF)wireless ments show that this PA offers 21.1 dBm saturated power and connectivity.Nowadays,CMOS RF building blocks,such as 18.4 dBm at 1dBCP to the load without thick-oxide MOS,the LNAs[1],mixers[2]and VCOs[31,have achieved good perfor- maximum PAE is measured to be 35.4%.With the adopted har- mance compared with other technologies,except for the PA be- monics termination,the output harmonics are also well con- cause CMOS technology suffers from some serious problems. trolled,as will be shown. Firstly,with a lossy substrate,a poor on-chip passive compo- nent such as an inductor may degrade the whole efficiency of 2.Basic theory of linear PA the PA.Meanwhile,substrate coupling may also cause some fatal problems to other blocks.Secondly,a low voltage supply The basic circuit of a linear PA is shown in Fig.1.Different scaled with technology limits the output power and linearity from a switch-mode PA,the power transistor MI is modeled dramatically.Critically,its low breakdown voltage is a catas- as a voltage-controlled current source,Lc is the RF choke for trophic factor for a PA's reliability if it tries to deliver a high feeding power to the drain and blocking RF signals,CB is a output level.In addition,the low mobility of CMOS results in through capacitor between the drain and output load,Ls with a much larger dimension for transistors,so its performance at Cp forms a LC tank at the working frequency for filtering the high frequency may be deteriorated because of a complicated unwanted harmonics,and RL represents the load of the PA.VB parasitic effect.Finally,the hot carrier effect also affects the performance and reliability.As a result,a higher integration for the on-chip PA becomes more challenging. Recently,most wireless standards would like to use non- constant envelope modulation schemes to enhance the data rate Matching and the system capacity.Taking the RFID system as an ex- Network ample,ASK is adopted to simplify the complexity and lower power consumption for passive tags that acquire energy from electromagnetic waves.As a result,a linear PA is still a good solution to deal with the variable envelope modulated signals However,one disadvantage of the linear PA is the efficiency, which is critical for use-time in practical applications.Another problem is how to achieve a high power and efficiency in deep sub-micro CMOS technology with a scaled supply. In this paper,a two-stage differential class-AB CMOS PA for a UHF RFID reader is presented.To overcome the reliabil- Fig.1.Basic topology of linear power amplifier. *Project supported by the Ministry of Science&Technology of China(No.2008BAI55B07). Corresponding author.Email:tanxi@fudan.edu.cn Received 21 April 2010,revised manuscript received 25 July 2010 C2010 Chinese Institute of Electronics 125005-1

Vol. 31, No. 12 Journal of Semiconductors December 2010 A 900 MHz, 21 dBm CMOS linear power amplifier with 35% PAE for RFID readers Han Kefeng(韩科锋), Cao Shengguo(曹圣国), Tan Xi(谈熙) Ž , Yan Na(闫娜), Wang Junyu(王俊宇), Tang Zhangwen(唐长文), and Min Hao(闵昊) (State Key Laboratory of ASIC & System, Fudan University, Shanghai 201203, China) Abstract: A two-stage differential linear power amplifier (PA) fabricated by 0.18 m CMOS technology is presented. An output matching and harmonic termination network is exploited to enhance the output power, efficiency and har￾monic performance. Measurements show that the designed PA reaches a saturated power of 21.1 dBm and the peak power added efficiency (PAE) is 35.4%, the power gain is 23.3 dB from a power supply of 1.8 V and the harmonics are well controlled. The total area with ESD protected PAD is 1:2 0:55 mm2 . System measurements also show that this power amplifier meets the design specifications and can be applied for RFID reader. Key words: CMOS; power amplifier; PAE; power matching; RFID; reader DOI: 10.1088/1674-4926/31/12/125005 EEACC: 1220 1. Introduction During the past several decades, CMOS has achieved great success for low cost and high yield, and it is also a popular choice of system on chip for radio frequency (RF) wireless connectivity. Nowadays, CMOS RF building blocks, such as LNAsŒ1, mixersŒ2 and VCOsŒ3, have achieved good perfor￾mance compared with other technologies, except for the PA be￾cause CMOS technology suffers from some serious problems. Firstly, with a lossy substrate, a poor on-chip passive compo￾nent such as an inductor may degrade the whole efficiency of the PA. Meanwhile, substrate coupling may also cause some fatal problems to other blocks. Secondly, a low voltage supply scaled with technology limits the output power and linearity dramatically. Critically, its low breakdown voltage is a catas￾trophic factor for a PA’s reliability if it tries to deliver a high output level. In addition, the low mobility of CMOS results in a much larger dimension for transistors, so its performance at high frequency may be deteriorated because of a complicated parasitic effect. Finally, the hot carrier effect also affects the performance and reliability. As a result, a higher integration for the on-chip PA becomes more challenging. Recently, most wireless standards would like to use non￾constant envelope modulation schemes to enhance the data rate and the system capacity. Taking the RFID system as an ex￾ample, ASK is adopted to simplify the complexity and lower power consumption for passive tags that acquire energy from electromagnetic waves. As a result, a linear PA is still a good solution to deal with the variable envelope modulated signals. However, one disadvantage of the linear PA is the efficiency, which is critical for use-time in practical applications. Another problem is how to achieve a high power and efficiency in deep sub-micro CMOS technology with a scaled supply. In this paper, a two-stage differential class-AB CMOS PA for a UHF RFID reader is presented. To overcome the reliabil￾ity problem, a cascode topology is employed. To gain a power of more than 20 dBm from a supply of 1.8 V, output matching for this differential PA is analyzed in detail and realized on a PCB board. To gain a higher efficiency and maintain relatively good linearity, the PA works in deep class-AB mode. Measure￾ments show that this PA offers 21.1 dBm saturated power and 18.4 dBm at 1dBCP to the load without thick-oxide MOS, the maximum PAE is measured to be 35.4%. With the adopted har￾monics termination, the output harmonics are also well con￾trolled, as will be shown. 2. Basic theory of linear PA The basic circuit of a linear PA is shown in Fig. 1. Different from a switch-mode PA, the power transistor M1 is modeled as a voltage-controlled current source, LC is the RF choke for feeding power to the drain and blocking RF signals, CB is a through capacitor between the drain and output load, LS with Cp forms a LC tank at the working frequency for filtering the unwanted harmonics, and RL represents the load of the PA. VB Fig. 1. Basic topology of linear power amplifier. * Project supported by the Ministry of Science & Technology of China (No. 2008BAI55B07). Ž Corresponding author. Email: tanxi@fudan.edu.cn Received 21 April 2010, revised manuscript received 25 July 2010 c 2010 Chinese Institute of Electronics 125005-1

J.Semicond.2010,31(12) Han Kefeng et al. PA type: B AB be divided by the coupling capacitor and the input parasitic ca- pacitor of the transistors,otherwise the whole gain will suffer 1.0e 100 from great loss. The first stage is biased at a relatively high overdrive level 邑0.8 PAE 90 in order to maintain a good linearity.In our test settings,the bias voltage is 0.74 V.For the second stage,the bias voltage is set to 0.53 V.close to its knee voltage but a bit more than 0.6 80 simulation,thus the second stage works at the mode of class- AB,closer to class-B for exchange of efficiency.It is worth 0.4 Current 0 mentioning that the 3rd order distortion of the transconductor fundamental (gm)is around zerol6l. 0.2 60 The output matching and harmonic control network is cre- ated using discrete components,LI series with Cl is for the 50 2nd order harmonic termination and L2 with C2 is for output 0.5 1.0152.02.3 power matching to maximize the output power,which will be Conduction angle (rad/s) discussed in particular next. Fig.2.Current fundamental and PAE in terms of conduction angle. 3.2.Differential versus single-ended topology Generally,on-chip circuits are differential to suppress the is the DC bias for M1.which determines the conduction angle, noise from the substrate or supply and reject common-mode thus PA is classified to class A,AB,B or C with corresponding interferences,and even order harmonics can be eliminated ide- angles.Each class has different performances,such as gain, ally.However.most antennas are single-ended.so differential efficiency,and linearity. to single-ended conversion is inevitable either on-chip or off- For simple analysis,the output current fundamental id chip.On-chip conversion needs passive or active balun.The and PAE of a linear PA can be expressed in terms of the con- former is a good choice with the cost of chip size,while the duction angle 0,given as latter one has to deal with large signal swing.As a result,ef- innd =(20-sin20) ficiency degradation is evident and the linearity is a practical (1) problem too.So an off-chip balun is applied in our scheme. 2π Differential topology is also expected to have an extra 28-sin28 6 dBm power than a single-ended one,and this is desirable es- PAE= 4(sin9-6cos9×100%. (2) pecially for CMOS technology when the supply voltage scales down.From another point of view,the output balun acts as a Here,the range of conduction angle0 is from0°toI80°， power combiner to the load and,as a result,the output power and Figure 2 depicts the results of Egs.(1)and(2)with re- increases. spect to 0.As shown in Fig.2,class-A has a maximum output fundamental current which implies a higher gain by sacrificing 3.3.Single-stage versus two-stage scheme the PAE,while class-C gains a PAE of 100%with a penalty of In our transmitter scheme,an active up-mixer uses a LC less output power.Of all the cases,class-AB is the most widely tank as its load,so when it is cascaded directly with power tran- used choice for a tradeoff between linearity and efficiency. sistors,the input capacitance of the PA will greatly affect the frequency response of the LC load.To drive the power stage, 3.Design of the two-stage CMOS PA an inter-stage amplifier is placed between the up-mixer and the PA as a buffer and this reduces the input parasitic capac- 3.1.Circuit design of linear PA itance.One disadvantage is that the driver stage will consume The overall schematic of the two-stage PA is shown in a fixed power from the supply and the efficiency will be de- Fig.3.The transistors MI and M2 are configured as the first graded compared to the single-stage scheme.The gain of the stage with dimension of 192 um/0.18 um.The second stage is driver should be small because in the transmitter path,the sig- similar and the dimensions are 2700 um/0.18 um.The on-chip nal level is much higher than a receiver,so linearity plays an inductor LLI paralleled with CLI is as LC load of first stage, important role for SNR.High gain is often provided by the last the inductor is of center tapped differential typel5]to save chip stage and low gain is for the baseband circuit and up-mixer.In area,and the power supply is fed via the center tap.Inter-stage this way,the transmitter can satisfy the stringent output spec- matching is done by RC high pass network,and the bias of the trum requirement,such as the adjacent channel power ratio second stage is given by the resistors.The outputs of the PA (ACPR)7.So,the high output level of the first stage to the last are connected to an off-chip balun,and the power is supplied stage will exceed the linear range of the PA and,consequently, by off-chip inductors LBI and LB2. the spectrum of output power will not satisfy the emission re- Due to the large dimension of the transistors,simulated in- quirements. put parasitic capacitance of M3a or M3b is up to 4 pF when the transistors are biased at about 0.5 V,so the inductance of L 3.4.Cascode versus common source should not be too large and capacitor CLI should be chosen The breakdown effect has become a key factor for the out- carefully with the post-layout simulation.The large coupling put power and reliability for designing a PA in deep sub-micro capacitance of Ce3 and Co4 is preferable because the signal will CMOS technology.To a traditional CS stage,a gate is most 125005-2

J. Semicond. 2010, 31(12) Han Kefeng et al. Fig. 2. Current fundamental and PAE in terms of conduction angle. is the DC bias for M1, which determines the conduction angle, thus PA is classified to class A, AB, B or C with corresponding angles. Each class has different performances, such as gain, efficiency, and linearity. For simple analysisŒ4, the output current fundamental ifund and PAE of a linear PA can be expressed in terms of the con￾duction angle , given as ifund D irf .2 ￾ sin 2 / 2 ; (1) PAE D 2 ￾ sin 2 4 .sin  ￾  cos  / 100%: (2) Here, the range of conduction angle  is from 0ı to 180ı , and Figure 2 depicts the results of Eqs. (1) and (2) with re￾spect to . As shown in Fig. 2, class-A has a maximum output fundamental current which implies a higher gain by sacrificing the PAE, while class-C gains a PAE of 100% with a penalty of less output power. Of all the cases, class-AB is the most widely used choice for a tradeoff between linearity and efficiency. 3. Design of the two-stage CMOS PA 3.1. Circuit design of linear PA The overall schematic of the two-stage PA is shown in Fig. 3. The transistors M1 and M2 are configured as the first stage with dimension of 192 m/0.18 m. The second stage is similar and the dimensions are 2700 m/0.18 m. The on-chip inductor LL1 paralleled with CL1 is as LC load of first stage, the inductor is of center tapped differential typeŒ5 to save chip area, and the power supply is fed via the center tap. Inter-stage matching is done by RC high pass network, and the bias of the second stage is given by the resistors. The outputs of the PA are connected to an off-chip balun, and the power is supplied by off-chip inductors LB1 and LB2. Due to the large dimension of the transistors, simulated in￾put parasitic capacitance of M3a or M3b is up to 4 pF when the transistors are biased at about 0.5 V, so the inductance of LL1 should not be too large and capacitor CL1 should be chosen carefully with the post-layout simulation. The large coupling capacitance of Cc3 and Cc4 is preferable because the signal will be divided by the coupling capacitor and the input parasitic ca￾pacitor of the transistors, otherwise the whole gain will suffer from great loss. The first stage is biased at a relatively high overdrive level in order to maintain a good linearity. In our test settings, the bias voltage is 0.74 V. For the second stage, the bias voltage is set to 0.53 V, close to its knee voltage but a bit more than simulation, thus the second stage works at the mode of class￾AB, closer to class-B for exchange of efficiency. It is worth mentioning that the 3rd order distortion of the transconductor (gm/ is around zeroŒ6 . The output matching and harmonic control network is cre￾ated using discrete components, L1 series with C1 is for the 2nd order harmonic termination and L2 with C2 is for output power matching to maximize the output power, which will be discussed in particular next. 3.2. Differential versus single-ended topology Generally, on-chip circuits are differential to suppress the noise from the substrate or supply and reject common-mode interferences, and even order harmonics can be eliminated ide￾ally. However, most antennas are single-ended, so differential to single-ended conversion is inevitable either on-chip or off￾chip. On-chip conversion needs passive or active balun. The former is a good choice with the cost of chip size, while the latter one has to deal with large signal swing. As a result, ef￾ficiency degradation is evident and the linearity is a practical problem too. So an off-chip balun is applied in our scheme. Differential topology is also expected to have an extra 6 dBm power than a single-ended one, and this is desirable es￾pecially for CMOS technology when the supply voltage scales down. From another point of view, the output balun acts as a power combiner to the load and, as a result, the output power increases. 3.3. Single-stage versus two-stage scheme In our transmitter scheme, an active up-mixer uses a LC tank as its load, so when it is cascaded directly with power tran￾sistors, the input capacitance of the PA will greatly affect the frequency response of the LC load. To drive the power stage, an inter-stage amplifier is placed between the up-mixer and the PA as a buffer and this reduces the input parasitic capac￾itance. One disadvantage is that the driver stage will consume a fixed power from the supply and the efficiency will be de￾graded compared to the single-stage scheme. The gain of the driver should be small because in the transmitter path, the sig￾nal level is much higher than a receiver, so linearity plays an important role for SNR. High gain is often provided by the last stage and low gain is for the baseband circuit and up-mixer. In this way, the transmitter can satisfy the stringent output spec￾trum requirement, such as the adjacent channel power ratio (ACPR)Œ7. So, the high output level of the first stage to the last stage will exceed the linear range of the PA and, consequently, the spectrum of output power will not satisfy the emission re￾quirements. 3.4. Cascode versus common source The breakdown effect has become a key factor for the out￾put power and reliability for designing a PA in deep sub-micro CMOS technology. To a traditional CS stage, a gate is most 125005-2

J.Semicond.2010,31(12) Han Kefeng et al. VDD(IV8) Off-Chip 1*stage 0000 2 stage M3 'a=0.74V a=0.53V Fig.3.Schematic of designed CMOS two-stage linear PA for RFID reader. 200 =0.70 Z=IK Z=Zon 150 0.65 Z 1/Z 2 0.60 Z=1 0.55 50 BIAS 0.5 1.5 1.82.0 (V) Fig.4.Relationship of load,output current and voltage likely to suffer from breakdown because it undergoes a large load is 12.92 dBm if a knee voltage of 0.4 V is considered anti-phase signal swing with the drain.If a cascode transistor for a single-ended circuit without any output matching method. is stacked on the CS transistor,the risk of breakdown can be Though the output level can be 6 dBm more for a differential reducedl).In this work,normal transistors are applied rather type,the power is still less than 20 dBm without any output than thick-oxide MOS because the maximum output swing at matching. the drain is 2Vpp for the linear type of PA.So,even in the worst Actually,when the dimension of a power transistor is cho- case,there is still enough margin to keep the cascode transis- sen,the maximum output power it can deliver to the load is tors from breakdown if the gate is biased close to 1.8 V for the determinedl]because the peak current of this power transistor cascode transistor[16]. is limited,so in theory the ultimate power(in dBm)that the PA Another issue is a stability problem which is very common can deliver is in the PA design.For the CS stage,a feedback resistor is often placed between the gate and the drain for the sake of stability PmaxldBm =101g VpDImax×1000 2 (3) In the cascode type PA,isolation can be improved by a cascode transistor without the feedback resistor.It should be noted that Here,Imax is the peak current of the power transistor.Actu- the reverse feed-through is also an important factor for insta- bility in the PA.The penalty of the cascode is that its linear- ally,Pmax in Eq.(3)cannot be reached because the maximum value of the output voltage and the current may not be achieved ity performance is inferior to the CS type-a difference of 2-5 dB can be found for the third-order inter-modulated component simultaneously.As shown in Fig.4,for a load of I k,the volt- age fast reaches a maximum value while the current is still far (IM3)for different bias to our simulation. away from its limit.In contrast,when the load is 19,the corre- 3.5.Output power matching sponding voltage is low enough when the current achieves its As the supply voltage scales down,it becomes more chal- peak.This case implies a conclusion that there is an optimal load in Eg.(3),and this optimized value can be expressed as lenging to realize a high output power PA on the chip.To a0.18 um CMOS technology,the typical supply voltage for normal VDD operation is 1.8 V,so the maximum power it can send to the 50 Zopt= Imax (4) 125005-3

J. Semicond. 2010, 31(12) Han Kefeng et al. Fig. 3. Schematic of designed CMOS two-stage linear PA for RFID reader. Fig. 4. Relationship of load, output current and voltage. likely to suffer from breakdown because it undergoes a large anti-phase signal swing with the drain. If a cascode transistor is stacked on the CS transistor, the risk of breakdown can be reducedŒ8. In this work, normal transistors are applied rather than thick-oxide MOS because the maximum output swing at the drain is 2VDD for the linear type of PA. So, even in the worst case, there is still enough margin to keep the cascode transis￾tors from breakdown if the gate is biased close to 1.8 V for the cascode transistorŒ16 . Another issue is a stability problem which is very common in the PA design. For the CS stage, a feedback resistor is often placed between the gate and the drain for the sake of stability. In the cascode type PA, isolation can be improved by a cascode transistor without the feedback resistor. It should be noted that the reverse feed-through is also an important factor for insta￾bility in the PA. The penalty of the cascode is that its linear￾ity performance is inferior to the CS type—a difference of 2–5 dB can be found for the third-order inter-modulated component (IM3) for different bias to our simulation. 3.5. Output power matching As the supply voltage scales down, it becomes more chal￾lenging to realize a high output power PA on the chip. To a 0.18 m CMOS technology, the typical supply voltage for normal operation is 1.8 V, so the maximum power it can send to the 50  load is 12.92 dBm if a knee voltage of 0.4 V is considered for a single-ended circuit without any output matching method. Though the output level can be 6 dBm more for a differential type, the power is still less than 20 dBm without any output matching. Actually, when the dimension of a power transistor is cho￾sen, the maximum output power it can deliver to the load is determinedŒ9 because the peak current of this power transistor is limited, so in theory the ultimate power (in dBm) that the PA can deliver is PmaxjdBm D 10 lg VDDImax 2 1000 : (3) Here, Imax is the peak current of the power transistor. Actu￾ally, Pmax in Eq. (3) cannot be reached because the maximum value of the output voltage and the current may not be achieved simultaneously. As shown in Fig. 4, for a load of 1 k, the volt￾age fast reaches a maximum value while the current is still far away from its limit. In contrast, when the load is 1, the corre￾sponding voltage is low enough when the current achieves its peak. This case implies a conclusion that there is an optimal load in Eq. (3), and this optimized value can be expressed as Zopt D VDD Imax : (4) 125005-3

J.Semicond.2010,31(12) Han Kefeng et al. 055 下子2附 Optimal Load Point Z=R+i Power Contour吨 (a) (b) Fig.5.Load-pull simulation and output matching methods The way to gain Px is called load-line matching or power matching.The practical value for optimal impedance (Zopt) Output Matching 900 MHz Path-A ideal model seen by the PA can be determined by complicated load-pull Path-B ideal model measurements,and the results may vary with the environmen- tal temperature,input power level,PCB parasitic parameters, Path-A: 7 device aging and so on. Path-A Figure 5(a)shows the simulated load pull power contours -10 Murata mode using spectra at an input of 0 dBm,and Figure 5(b)shows the Path-B matching methods to the optimal point.For a linear PA design, Murata mode the optimal point is often located in the shadowed circular re- Path-B: gion.So,generally speaking,Zopt is lower than the load RL. This work Thus for a power matching PA,the small signal gain will drop Murata model compared to a PA that drives the load directly,while in large signal condition,the output power will be improved and,ac- 3 cordingly,the efficiency is enhanced dramatically. 0.1 0.5 Frequency (GHz) Figure 6 shows the two types of output matching network and the simulated frequency responses for each case in Agi- Fig.6.Two types of matching network of the PA. lent ADS by an ideal or Murata component model for match- ing.Path B is adopted here to suppress more of the harmon- 1.20mm ics or spurious components at higher frequency.A new branch for 2nd order termination is inserted between the output balun and the matching network for two reasons.Firstly,the 2nd or- der distortion in gm is maximized when 3rd order distortion dStage is set to be zero intentionally.As a result,due to mismatch in differential outputs,the 2nd order component may exceed 3rd order harmonics in practice.Secondly,the capacitance of Cgd couples the 2nd order harmonics to the input,and 3rd order harmonics will be regenerated by fundamental and fed back 2nd order harmonicsl6).With the output matching and har- monic control network,the output power and harmonics will Fig.7.Die micrograph of the designed two-stage CMOS PA. be greatly improved,as will be shown in the section of mea- surements.The PAE will also be improved,together with more output power to the load,according to 4.Chip implementation and experimental results The chip was fabricated by SMIC in a 0.18 um CMOS PAE=Pou-Pin x 100%. PDc (5) mixed-signal 1P6M process.Its micrograph is shown in Fig.7. Multi-pads were applied for differential outputs and ground to where Pout and Pin refer to the output and input power respec- minimize the inductance of the bond-wire. tively,and Ppc means the total power of the PA driven from Input matching was done by an off-chip LC type network the supply. (not shown in Fig.3)and a balun(Mini-circuits:TC4-14G2+) 125005-4

J. Semicond. 2010, 31(12) Han Kefeng et al. Fig. 5. Load-pull simulation and output matching methods. The way to gain Pmax is called load-line matching or power matching. The practical value for optimal impedance (Zopt/ seen by the PA can be determined by complicated load-pull measurements, and the results may vary with the environmen￾tal temperature, input power level, PCB parasitic parameters, device aging and so on. Figure 5(a) shows the simulated load pull power contours using spectra at an input of 0 dBm, and Figure 5(b) shows the matching methods to the optimal point. For a linear PA design, the optimal point is often located in the shadowed circular re￾gion. So, generally speaking, Zopt is lower than the load RL. Thus for a power matching PA, the small signal gain will drop compared to a PA that drives the load directly, while in large signal condition, the output power will be improved and, ac￾cordingly, the efficiency is enhanced dramatically. Figure 6 shows the two types of output matching network and the simulated frequency responses for each case in Agi￾lent ADS by an ideal or Murata component model for match￾ing. Path B is adopted here to suppress more of the harmon￾ics or spurious components at higher frequency. A new branch for 2nd order termination is inserted between the output balun and the matching network for two reasons. Firstly, the 2nd or￾der distortion in gm is maximized when 3rd order distortion is set to be zero intentionally. As a result, due to mismatch in differential outputs, the 2nd order component may exceed 3rd order harmonics in practice. Secondly, the capacitance of Cgd couples the 2nd order harmonics to the input, and 3rd order harmonics will be regenerated by fundamental and fed back 2nd order harmonicsŒ6. With the output matching and har￾monic control network, the output power and harmonics will be greatly improved, as will be shown in the section of mea￾surements. The PAE will also be improved, together with more output power to the load, according to PAE D Pout ￾ Pin PDC 100%; (5) where Pout and Pin refer to the output and input power respec￾tively, and PDC means the total power of the PA driven from the supply. Fig. 6. Two types of matching network of the PA. Fig. 7. Die micrograph of the designed two-stage CMOS PA. 4. Chip implementation and experimental results The chip was fabricated by SMIC in a 0.18 m CMOS mixed-signal 1P6M process. Its micrograph is shown in Fig. 7. Multi-pads were applied for differential outputs and ground to minimize the inductance of the bond-wire. Input matching was done by an off-chip LC type network (not shown in Fig. 3) and a balun (Mini-circuits: TC4-14G2+) 125005-4

J.Semicond.2010,31(12) Han Kefeng et al. 25 30 Output 1dBCP 18.4 dBm 200 25 15 Power gain (ap) -20 0 power Without output 10 40 matching network 5 -60 Measured 900 MHz -80 -30 25 -20-15-10-5 0 0.5 0.91.0 5 Frequency(GHz) Input power (dBm) Fig.9.Power gain and output power versus input power. Fig.8.Measured S-parameter of the PA Mkr4 3.600 GHz was used for single-to-differential conversion.This balun with Ref 25 dBm Atten 40 dB 42.545dBm a typical loss of about 0.85 dB at 900 MHz was also applied Samp 10g to the outputs and the maximum RF power provided was 24 10 dB/ dBm,which was enough for this test. Two independent LDOs(NS:LM317)were used for the 3.3dB test PCB:one was for voltage bias and the other was a power 1Z.8&B .0dB supply for the PA.With 5 V input,the LDOs offered a 1.8 V supply and a maximum 1.5 A current to the load.Decoupling LgAv capacitors were placed carefully on the PCB to avoid instabil- Center 2,400 GHz Span 4 GHz ity,which is very common and hazardous in designing a PA. ·Res BW300kHz VBW 300 kHz Sweep 169.5 ms (601 pts) I-V curves of the two stages were firstly plotted by mea- suring the gate voltages and corresponding currents,then bias Fig.10.Measured harmonics with/without matching and harmonics voltages for the PA could be set according to the I-V curve. control network. The DC current of this PA was 35 mA after de-embedding the currents from the LDOs. 40 Scattering parameters were measured by an Agilent E5071B VNA from 0.5 to 2.5 GHz,as shown in Fig.8.Good 35H PAE 1dBCP=29.2% input matching(S1)was achieved at a frequency of 900 MHz and the power gain(S21)reached a peak value of 22 dB around PAE@peak =35.4% the desired frequency.S12 was about-38 dB,so the PA pro- 25-PAE backoff 4dB=20.0% vided good isolation and hence improved the stability. PAE@peak=17.5% 20 An Agilent E4440A PSA was used for power,gain and har- monics measurements of the device under test(DUT).In Fig.9. 15 the power gain is measured to be 23.3 dB at 900 MHz and the 10 saturated power is 21.1 dBm after de-embedding 1.3 dBm loss Without output of balun,PCB runner and test line to the instrument.The output matching network power at 1dBCP is up to 18.4 dBm and the corresponding in- 25 put power is about-4 dBm.The power and gain curves are also -20-15-10-505 given.Although the PA with a matching network suffers from a Input power (dBm) little gain loss for small signal inputs,the saturated power gains Fig.11.Measured power added efficiency of the PA about 3.7 dBm improvement.The harmonics(at least up to 4th order)are less than-50 dBc at the output power of 8.88 dBm (without calibrating the output loss)and the harmonics are still with the matching network now adopted in this work. below-40 dBc at their 1dBCP with the 2nd order termination. The IM3 and IM5 components shown in Fig.12 were mea- As shown in Fig.10,the harmonics are greatly improved with sured by a two-tone test with frequencies of 899 MHz and the output matching and 2nd order termination network in this 901 MHz.To make comparisons,the adjacent channel power work. ratio(ACPRI)and alternative channel power ratio (ACPR2) The maximum PAE is measured to be 35.4%and it drops are also given in Fig.12.The ACPR was measured by a 160- to 29.2%at 1dBCP,a little lower than its peak value,as shown kHz PIE DSB-ASK signal,which was generated by a vector in Fig.11.When back off 4 dB from 1 dBCP,the PAE is still signal generator(Agilent PSG 8267D).As shown in Fig.13, up to 20.0%.For the case of no output matching network,the the spectrum which meets the emission mask requirement de- peak PAE is only 17.5%,so the efficiency is greatly improved fined by EPC globall is measured by an input of-14 dBm 125005-5

J. Semicond. 2010, 31(12) Han Kefeng et al. Fig. 8. Measured S-parameter of the PA. was used for single-to-differential conversion. This balun with a typical loss of about 0.85 dB at 900 MHz was also applied to the outputs and the maximum RF power provided was 24 dBm, which was enough for this test. Two independent LDOs (NS: LM317) were used for the test PCB: one was for voltage bias and the other was a power supply for the PA. With 5 V input, the LDOs offered a 1.8 V supply and a maximum 1.5 A current to the load. Decoupling capacitors were placed carefully on the PCB to avoid instabil￾ity, which is very common and hazardous in designing a PA. I –V curves of the two stages were firstly plotted by mea￾suring the gate voltages and corresponding currents, then bias voltages for the PA could be set according to the I –V curve. The DC current of this PA was 35 mA after de-embedding the currents from the LDOs. Scattering parameters were measured by an Agilent E5071B VNA from 0.5 to 2.5 GHz, as shown in Fig. 8. Good input matching (S11) was achieved at a frequency of 900 MHz and the power gain (S21) reached a peak value of 22 dB around the desired frequency. S12 was about –38 dB, so the PA pro￾vided good isolation and hence improved the stability. An Agilent E4440A PSA was used for power, gain and har￾monics measurements of the device under test (DUT). In Fig. 9, the power gain is measured to be 23.3 dB at 900 MHz and the saturated power is 21.1 dBm after de-embedding 1.3 dBm loss of balun, PCB runner and test line to the instrument. The output power at 1dBCP is up to 18.4 dBm and the corresponding in￾put power is about –4 dBm. The power and gain curves are also given. Although the PA with a matching network suffers from a little gain loss for small signal inputs, the saturated power gains about 3.7 dBm improvement. The harmonics (at least up to 4th order) are less than –50 dBc at the output power of 8.88 dBm (without calibrating the output loss) and the harmonics are still below –40 dBc at their 1dBCP with the 2nd order termination. As shown in Fig. 10, the harmonics are greatly improved with the output matching and 2nd order termination network in this work. The maximum PAE is measured to be 35.4% and it drops to 29.2% at 1dBCP, a little lower than its peak value, as shown in Fig. 11. When back off 4 dB from 1 dBCP, the PAE is still up to 20.0%. For the case of no output matching network, the peak PAE is only 17.5%, so the efficiency is greatly improved Fig. 9. Power gain and output power versus input power. Fig. 10. Measured harmonics with/without matching and harmonics control network. Fig. 11. Measured power added efficiency of the PA. with the matching network now adopted in this work. The IM3 and IM5 components shown in Fig. 12 were mea￾sured by a two-tone test with frequencies of 899 MHz and 901 MHz. To make comparisons, the adjacent channel power ratio (ACPR1) and alternative channel power ratio (ACPR2) are also given in Fig. 12. The ACPR was measured by a 160- kHz PIE DSB-ASK signal, which was generated by a vector signal generator (Agilent PSG 8267D). As shown in Fig. 13, the spectrum which meets the emission mask requirement de￾fined by EPC globalŒ14 is measured by an input of –14 dBm 125005-5

J.Semicond.2010,31(12) Han Kefeng et al. Table 1.Performances summary and comparisons with other works Parameter Ref.[6] Ref.[10] Ref.[11] Ref.[12] Ref.[13] This work CMOS process 0.18 um Cu 0.25um 0.18um 0.254m 0.54m 0.184m Power supply (V) 2.5 2.5 2.4 2.5 3.3 1.8 Frequency (GHz) 2.45 0.9 2.4 2.45 1.75 0.9 PA mode Class-AB Class-AB Self-biased Class-AB Class-AB Class-AB Power gain(dB) 17.5 14.5 38 11.2 23.9 23.3 Power (dBm)Saturated N/A N/A 23.5 N/A N/A 21.1 1 dBCP 20.5 27 20.5 20 24 18.4 PAE(%) Saturated N/A N/A 45 0 33 35.4 1 dBCP 37 28 17 28 2 29.2 Back-off 28(4dB) 7(5dB) 7(4dB) (4dB)* 17(5dB) 20(4dB) PA stages 2 2 2 2 2 2 Size(mm2) 1.34 4.4* 0.46 0.77 3.2 0.66 Estimated from the paper. -10 Ch Freq 908Hz Trig Free Adj Channel Power Averages:100 20 ACPRI requirement IM3 Average 100 -304 ACPRI Ref 15 dBm Atten 30 dB WAvg Log -40 dB/ IM5 ACPR2 -50 ACPR2 requirement -6 Center 900.080 0 MHz pan 2.88 MHz Res BW 3 kHz VBW 30 kHz Sweep934ms(681pts】 RMS Results Fraa off of BLI HBc Lovar Bc Upper dB Carrler Power 200 -20 15 .10 5 B.22 dBn/ Input power(dBm） 320.80BkHz 288 MH- -6957 Fig.12.Measured inter-modulated components and ACPR Fig.13.Measured spectrum for a 160 kHz DSB-ASK input at 8.22 and the RMS channel power is 8.22 dBm without calibrating dBm output. the loss. This PA is also integrated into the RFID reader transceiver low-noise amplifier exploiting thermal noise canceling.IEEE J chip[5].According to the system measurements with a digital Solid-State Circuits,2004,39(2):275 baseband,the reader with an integrated PA can read the infor- [2]Zhuo W.Embabi S,de Gyvez J P.et al.Using capacitive mation reflected by the passive tag. cross-coupling technique in RF low noise amplifier and down- conversion mixer design.IEEE Proc ESSCIRC,2000:77 5.Conclusion [3]Chen S F,Lee Y B,Sun C H,et al.A 65 nm CMOS low-noise direct-conversion transmitter with carrier leakage calibration for A linear PA used for a RFID reader was implemented in low-band EDGE application.IEEE Proc RFIC,2009:193 CMOS 0.18-um technology.With an output matching network [4]Lee T H.The design of CMOS radio-frequency integrated cir- and a harmonic control strategy,measurements show that this cuits.2nd ed.Beijing:Publishing House of Electronics Industry, 2005 PA achieves a saturated power of 21.1 dBm,a power gain of 23.3 dB and a maximum PAE of 35.4%from a 1.8 V supply [5]Lu Lei,Zhou Feng,Tang Zhangwen,et al.Equivalent model and at 900 MHz.The harmonics are measured to be less than-40 parameter extraction of center-tapped differential inductors.Chi- nese Journal of Semiconductors,2006,27(12):2150 dBc at its 1 dBCP. [6]Kang J,Yoon J,Min K,et al.A highly linear and efficient dif Table I summarizes the performances of this work,and ferential CMOS power amplifier with harmonic control.IEEE J makes comparisons with similar work.From this table,it can Solid-State Circuits,2006,41(6):1314 be seen that the performances of this work are comparable with [7]Gu Q.RF system design of transceivers for wireless communica- other work from a 1.8 V supply.Other work with higher power tions.New York:Springer,2005 supplies do not emphasize the matching network,so the output [8]Jeong J,Pornpromlikit S,Asbeck P M,et al.A 20 dBm linear power and efficiency may suffer from a little degradation. RF power amplifier using stacked silicon-on-sapphire MOSFETs. IEEE Microw Wireless Compon Lett,2006,16(12):684 References [9]Cripps S C.RF power amplifiers for wireless communications. 2nd ed.Norwood:Artech,2006 [1]Bruccoleri F,Klumperink E A M,Nauta B.Wide-band CMOS [10]Han J,Kim Y,Park C,et al.A fully-integrated 900-MHz CMOS 125005-6

J. Semicond. 2010, 31(12) Han Kefeng et al. Table 1. Performances summary and comparisons with other works. Parameter Ref. [6] Ref. [10] Ref. [11] Ref. [12] Ref. [13] This work CMOS process 0.18 m Cu 0.25 m 0.18 m 0.25 m 0.5 m 0.18 m Power supply (V) 2.5 2.5 2.4 2.5 3.3 1.8 Frequency (GHz) 2.45 0.9 2.4 2.45 1.75 0.9 PA mode Class-AB Class-AB Self-biased Class-AB Class-AB Class-AB Power gain (dB) 17.5 14.5 38 11.2 23.9 23.3 Power (dBm) Saturated N/A N/A 23.5 N/A N/A 21.1 1 dBCP 20.5 27 20.5 20 24 18.4 PAE (%) Saturated N/A N/A 45 30 33 35.4 1 dBCP 37 28 17 28 29 29.2 Back-off 28 (4 dB) 17 (5 dB) 7 (4 dB) 13 (4 dB) 17 (5 dB) 20 (4 dB) PA stages 2 2 2 2 2 2 Size (mm2 / 1.34 4.4 0.46 0.77 3.2 0.66 * Estimated from the paper. Fig. 12. Measured inter-modulated components and ACPR. and the RMS channel power is 8.22 dBm without calibrating the loss. This PA is also integrated into the RFID reader transceiver chipŒ15. According to the system measurements with a digital baseband, the reader with an integrated PA can read the infor￾mation reflected by the passive tag. 5. Conclusion A linear PA used for a RFID reader was implemented in CMOS 0.18-m technology. With an output matching network and a harmonic control strategy, measurements show that this PA achieves a saturated power of 21.1 dBm, a power gain of 23.3 dB and a maximum PAE of 35.4% from a 1.8 V supply at 900 MHz. The harmonics are measured to be less than ￾40 dBc at its 1 dBCP. Table 1 summarizes the performances of this work, and makes comparisons with similar work. From this table, it can be seen that the performances of this work are comparable with other work from a 1.8 V supply. Other work with higher power supplies do not emphasize the matching network, so the output power and efficiency may suffer from a little degradation. References [1] Bruccoleri F, Klumperink E A M, Nauta B. Wide-band CMOS Fig. 13. Measured spectrum for a 160 kHz DSB-ASK input at 8.22 dBm output. low-noise amplifier exploiting thermal noise canceling. IEEE J Solid-State Circuits, 2004, 39(2): 275 [2] Zhuo W, Embabi S, de Gyvez J P, et al. Using capacitive cross-coupling technique in RF low noise amplifier and down￾conversion mixer design. IEEE Proc ESSCIRC, 2000: 77 [3] Chen S F, Lee Y B, Sun C H, et al. A 65 nm CMOS low-noise direct-conversion transmitter with carrier leakage calibration for low-band EDGE application. IEEE Proc RFIC, 2009: 193 [4] Lee T H. The design of CMOS radio-frequency integrated cir￾cuits. 2nd ed. Beijing: Publishing House of Electronics Industry, 2005 [5] Lu Lei, Zhou Feng, Tang Zhangwen, et al. Equivalent model and parameter extraction of center-tapped differential inductors. Chi￾nese Journal of Semiconductors, 2006, 27(12): 2150 [6] Kang J, Yoon J, Min K, et al. A highly linear and efficient dif￾ferential CMOS power amplifier with harmonic control. IEEE J Solid-State Circuits, 2006, 41(6): 1314 [7] Gu Q. RF system design of transceivers for wireless communica￾tions. New York: Springer, 2005 [8] Jeong J, Pornpromlikit S, Asbeck P M, et al. A 20 dBm linear RF power amplifier using stacked silicon-on-sapphire MOSFETs. IEEE Microw Wireless Compon Lett, 2006, 16(12): 684 [9] Cripps S C. RF power amplifiers for wireless communications. 2nd ed. Norwood: Artech, 2006 [10] Han J, Kim Y, Park C, et al. A fully-integrated 900-MHz CMOS 125005-6

J.Semicond.2010,31(12) Han Kefeng et al. power amplifier for mobile RFID reader applications.IEEE Proc AB power amplifiers.IEEE J Solid-State Circuits,2004,39(11): RFIC.2006 1927 [11]Sowlati T,Leenaerts DM W.A 2.4-GHz 0.18 um CMOS self-[14]EPC UHF radio frequency identity protocols:Class 1 Generation biased cascode power amplifier.IEEE J Solid-State Circuits, 2 UHF RFID Ver.1.2.0,EPC global,2007 2003.38(8):1318 [15]Tan Xi,Liu Yuan,Lu Lei,et al.A 1.8 V CMOS direct conversion [12]Yen CC,Chuang H R.A 0.25-um 20-dBm 2.4-GHz CMOS receiver for 900 MHz RFID reader chip.Journal of Semiconduc- power amplifier with an integrated diode linearizer.IEEE Microw tors,2008,29(9):1734 Wireless Compon Lett,2003,13(2):45 [16]Si WW,Mehta S,Samavati H,et al.A 1.9-GHz single-chip [13]Wang C,Vaidyanathan M,Larson L E.A capacitance- CMOS PHS cellphone.IEEE J Solid-State Circuits,2006,41(12): compensation technique for improved linearity in CMOS class- 2737 125005-7

J. Semicond. 2010, 31(12) Han Kefeng et al. power amplifier for mobile RFID reader applications. IEEE Proc RFIC, 2006 [11] Sowlati T, Leenaerts D M W. A 2.4-GHz 0.18 m CMOS self￾biased cascode power amplifier. IEEE J Solid-State Circuits, 2003, 38(8): 1318 [12] Yen C C, Chuang H R. A 0.25-m 20-dBm 2.4-GHz CMOS power amplifier with an integrated diode linearizer. IEEE Microw Wireless Compon Lett, 2003, 13(2): 45 [13] Wang C, Vaidyanathan M, Larson L E. A capacitance￾compensation technique for improved linearity in CMOS class￾AB power amplifiers. IEEE J Solid-State Circuits, 2004, 39(11): 1927 [14] EPC UHF radio frequency identity protocols: Class 1 Generation 2 UHF RFID Ver. 1.2.0, EPC global, 2007 [15] Tan Xi, Liu Yuan, Lu Lei, et al. A 1.8 V CMOS direct conversion receiver for 900 MHz RFID reader chip. Journal of Semiconduc￾tors, 2008, 29(9): 1734 [16] Si W W, Mehta S, Samavati H, et al. A 1.9-GHz single-chip CMOS PHS cellphone. IEEE J Solid-State Circuits, 2006, 41(12): 2737 125005-7